HIGHLY LINEAR EMBEDDED FILTERING PASSIVE MIXER

• US 20090252252A1
• Filed: 08/15/2008
• Published: 10/08/2009
• Est. Priority Date: 04/07/2008
• Status: Active Grant

• Current Assignee: Qualcomm, Inc.
• Original Assignee: Qualcomm, Inc.
• Inventors: Kim, Namsoo, Gudem, Prasad S., Aparin, Vladimir
• Granted Patent: US 8,331,897 B2
• Time in Patent Office: Days
• US Class Current: 375/268
• CPC Class Codes: H03D 7/1466 Passive mixer arrangements
• CPC Class Codes: H03D 7/163 the local oscillations of at least two of the frequency changers being derived from a single oscillator
• CPC Class Codes: H03D 7/166 using two or more quadrature frequency translation stages

Abstract

A communication channel has a highly linear switched current mixer that incorporates passive filtering (e.g., low pass, notch) for improved transmitting (Tx) and receiving (Rx) with adding external filtering components. A high IIP₂ (input referenced second order intercept point) of the receiver at the Tx offset is essential to avoid corrupting the system'"'"'s sensitivity performance, and a high triple beat (TB) is required to avoid sensitivity degradation due to transmitter leakage. Thanks to the embedded filtering in the mixer and the active post-distortion (APD) method in a low noise amplifier (LNA), the required high linearity is achieved with low noise figure and power consumption, overcoming transmitter power leakage without the use of a SAW (surface acoustic wave) filter.

Claims

1. An integrated circuit for highly linear mixing of a radio frequency (RF) signal in a mixer circuitry, the mixer circuitry comprising:

a local oscillator;

a first switching stage for frequency converting an RF signal switched by the local oscillator;

passive filter circuitry electrically connected to output of the first stage; and

a second switching stage switched by the local oscillator and electrically connected to the output of the passive filter circuitry to convert charge into a current signal.

2. The integrated circuit of claim 1, further comprising active post-distortion (APD) method in a low noise amplifier (LNA) electrically connected to receive the RF signal for amplification before the mixer circuitry to form a receiver.

3. The integrated circuit of claim 1, wherein the passive filter circuitry of claim 2 further comprises a low pass filter.

4. The integrated circuit of claim 3, further comprising a transimpedance amplifier electrically connected to outputs of the mixer circuitry.

5. The integrated circuit of claim 3, further comprising a plurality of mixer circuitry, each tuned to a separate discrete channel for separately tuning an RF multi-channel media communication signal.

6. The integrated circuit of claim 1, further comprising:

a digital-to-analog converter (DAC) for converting a received digital signal into an analog signal;

a baseband filter electrically connected to the DAC for removing noise caused by the conversion and electrically connected to the mixer circuitry for providing the RF signal; and

a power amplifier electrically connected to the mixer circuitry for amplifying the output from the mixer circuitry for transmission,

wherein the passive filter circuitry of the mixer circuitry comprises a notch filter at a receiver (RX) band.

7. The integrated circuit of claim 1, wherein the local oscillator (LO) signal produces opposite phases V_LOIP and V_LOIM, the embedded passive filter comprises a series Resistor-Capacitor (RC) filter that determines filtering bandwidth, wherein radio frequency (RF) current i_RFp, i_RFm flows into the first set switch for down-conversion as a nominal passive mixer having a current and a voltage across when the positive LO signal (V_LOIP) is high as expressed by

$\begin{array}{c}{i}_A\end{array}$

( t ) ≈

g m , LNA ·

A rf ·



- 1 / τ

·

2 

f LO ·

[ cos 

( ω

rf 

t ) + 2 3 

π

·

cos 

( 2 

ω

rf 

t ) - 2 15 

π

·

cos 

( 4 

ω

rf 

t ) 



…

] ·

LO 

( t ) 





and v C m = 1 C m ·

∫

i A 

( t ) 



t ·

[ 1 + ( f offset / f 3 



dB ) 2 ] - 1 2 ,

where, τ

=R_mC_m, f_{3 dB}=(2π

)<sup>−

1</sup>, A_rf is amplitude of RF input voltage, and g_m,LNA is the transconductance of the LNA.

8. A method for highly linear mixing of a radio frequency (RF) signal, comprising:

receiving an RF signal at an integrated circuit mixer;

locally oscillating a first switching stage to perform frequency conversion of the RF signal;

passing the frequency converted signal through an integral passive filter to suppress a noise component; and

passing the passively filtered signal through a locally oscillated second switching stage to output a mixed signal.

9. The method of claim 8, further comprising passing the frequency converted signal through a low pass integral passive filter to suppress transmitter leakage.

10. The method of claim 9, further comprising performing frequency conversion at a first integrated circuit mixer of a plurality integrated circuit mixers of a tuner, each integrated circuit mixer tuned to a unique, discrete channel of the RF signal.

11. The method of claim 8, further comprising passing the frequency converted signal through a notch integral passive filter to suppress transmitter noise at a receiver band for transmission of a signal after mixing.

12. The method of claim 8, wherein the local oscillator (LO) signal produces opposite phases V_LOIP and V_LOIM, the embedded passive filter comprises a series Resistor-Capacitor (RC) filter that determines filtering bandwidth, wherein radio frequency (RF) current i_RFp, i_RFm flows into the first set switch for down-conversion as a nominal passive mixer having a current and a voltage across when the positive LO signal (V_LOIP) is high as expressed by

$\begin{array}{c}{i}_A\end{array}$

( t ) ≈

g m , LNA ·

A rf ·



- 1 / τ

·

2 

f LO ·

[ cos 

( ω

rf 

t ) + 2 3 

π

·

cos 

( 2 

ω

rf 

t ) - 2 15 

π

·

cos 

( 4 

ω

rf 

t ) 



…

] ·

LO 

( t ) 



and v C m = 1 C m ·

∫

i A 

( t ) 



t ·

[ 1 + ( f offset / f 3 



dB ) 2 ] - 1 2 ,

where, τ

=R_mC_m, f_{3 dB}=(2π

)<sup>−

1</sup>, A_rf is amplitude of RF input voltage, and g_m,LNA is the transconductance of the LNA.

13. An apparatus for highly linear mixing of a radio frequency (RF) signal, comprising:

means for receiving an RF signal at an integrated circuit mixer;

means for locally oscillating a first switching stage to perform frequency conversion of the RF signal;

means for passing the frequency converted signal through an integral passive filter to suppress a noise component; and

means for passing the passively filtered signal through a locally oscillated second switching stage to output a mixed signal.

14. The apparatus of claim 13, further comprising means for passing the frequency converted signal through a low pass integral passive filter to suppress transmitter leakage.

15. The apparatus of claim 14, further comprising means for performing frequency conversion at a first integrated circuit mixer of a plurality integrated circuit mixers of a tuner, each integrated circuit mixer tuned to a unique, discrete channel of the RF signal.

16. The apparatus of claim 13, further comprising means for passing the frequency converted signal through a notch integral passive filter to suppress transmitter noise at a receiver band for transmission of a signal after mixing.

17. The apparatus of claim 13, wherein the local oscillator (LO) signal produces opposite phases V_LOIP and V_LOIM, the embedded passive filter comprises a series Resistor-Capacitor (RC) filter that determines filtering bandwidth, wherein radio frequency (RF) current i_RFp, i_RFm flows into the first set switch for down-conversion as a nominal passive mixer having a current and a voltage across when the positive LO signal (V_LOIP) is high as expressed by

$\begin{array}{c}{i}_A\end{array}$

( t ) ≈

g m , LNA ·

A rf ·



- 1 / τ

·

2 

f LO ·

[ cos 

( ω

rf 

t ) + 2 3 

π

·

cos 

( 2 

ω

rf 

t ) - 2 15 

π

·

cos 

( 4 

ω

rf 

t ) 



…

] ·

LO 

( t ) 





and v C m = 1 C m ·

∫

i A 

( t ) 



t ·

[ 1 + ( f offset / f 3 



dB ) 2 ] - 1 2 ,

where, τ

=R_mC_m, f_{3 dB}=(2π

)<sup>−

1</sup>, A_rf is amplitude of RF input voltage, and g_m,LNA is the transconductance of the LNA.