VARIABLE DC LINK CONVERTER AND TRANSFORMER FOR WIDE OUTPUT VOLTAGE RANGE APPLICATIONS
1. A power converter, comprising:
- a first converter stage configured to convert power from a power source to power at an intermediate link voltage;
a second converter stage configured to convert the power at the intermediate link voltage to power for charging a battery; and
a control system comprising;
an intermediate link voltage regulation control loop configured, in a first mode of operation, to regulate the intermediate link voltage through the first converter stage based on a voltage of the battery; and
a ripple regulation control loop configured to sense a charging current for the battery and regulate a gain of the second converter stage based on the charging current to reduce ripple in the charging current.
A variable direct current (DC) link power converter is described. In one example, the power converter includes a first converter stage configured to convert power from a power source to power at an intermediate link voltage and a second converter stage configured to convert the power at the intermediate link voltage to power for charging a battery. The power converter further includes a control system having an intermediate link voltage regulation control loop configured, in a first mode of operation, to regulate the intermediate link voltage through the first converter stage based on a voltage of the battery, and a ripple regulation control loop configured to sense a charging current for the battery and regulate a gain of the second converter stage based on the charging current to reduce ripple in the charging current. A new configuration of transformer suitable for use with the power converter is also described.
|Voltage converting system and method of using the same|
Patent #US 10,476,395 B2
Current AssigneeFuturewei Technologies Incorporated
Sponsoring EntityFuturewei Technologies Incorporated
|Passive circuit and power converter|
Patent #US 10,608,521 B2
Current AssigneeDelta Electronics Incorporated
Sponsoring EntityDelta Electronics Incorporated
- 1. A power converter, comprising:
a first converter stage configured to convert power from a power source to power at an intermediate link voltage; a second converter stage configured to convert the power at the intermediate link voltage to power for charging a battery; and a control system comprising; an intermediate link voltage regulation control loop configured, in a first mode of operation, to regulate the intermediate link voltage through the first converter stage based on a voltage of the battery; and a ripple regulation control loop configured to sense a charging current for the battery and regulate a gain of the second converter stage based on the charging current to reduce ripple in the charging current.
- View Dependent Claims (2, 3, 4, 5, 6, 7, 8, 9, 10, 11)
- 12. A power converter, comprising:
a first converter stage and a second converter stage configured to convert power using an intermediate link voltage; and a control system comprising; an intermediate link voltage regulation control loop configured to regulate the intermediate link voltage through at least one of the first converter stage or the second converter stage; and a ripple regulation control loop configured to regulate a gain of the first converter stage or the second converter stage to reduce ripple in the current output by the power converter.
- View Dependent Claims (13, 14, 15, 16, 17)
- 18. The power converter of claim 18, wherein a number of windings of the primary winding on a first of the outer posts of the E-shaped core is different than a number of windings of the primary winding on a second of the outer posts of the E-shaped core.
This application claims the benefit of U.S. Provisional Application No. 62/395,134, filed Sep. 15, 2016, the entire contents of which is hereby incorporated herein by reference.
Power conversion is related to the conversion of electric power or energy from one form to another. Power conversion can involve converting between alternating current (AC) and direct current (DC) forms of energy, changing the voltage, current, or frequency of energy, or changing some other aspect of energy from one form to another. In that context, a power converter is an electrical or electro-mechanical device for converting electrical energy. A transformer is one example of a power converter, although more complicated systems, including complex arrangements of switching transistors, transformers, and control loops, can be used.
Many aspects of the present disclosure can be better understood with reference to the following drawings. The components in the drawings are not necessarily drawn to scale, with emphasis instead being placed upon clearly illustrating the principles of the disclosure. In the drawings, like reference numerals designate corresponding parts throughout the several views.
One challenge for AC/DC power converters is to accommodate the demands of lightweight, compact, efficient, and low cost systems. It can be difficult because, for some applications, such as power converters used for battery chargers, the output voltage range of such converters needs to be relatively wide. For example, from the fully charged to fully depleted range, a 2:1 ratio or greater can be expected.
In that context,
With conventional two stage battery chargers, the DC-link voltage between the first AC/DC stage and the second DC/DC stage is usually constant. As an example,
The drawback of battery charger circuit 20 is obvious, as the gain range (Vo/Vin) of the second DC/DC stage is wide.
For DC/DC converters, whether it is resonant type (e.g., LLC, CLLC, SRC, etc.), phase shift type (e.g., dual active bridge or phase shift full bridge), or pulse width modulation (PWM) type (e.g., forward or flyback), each has only one point that can achieve the highest efficiency with a fixed design. As the output requirements change, working conditions will drift the converter from its optimized point and efficiency will suffer, as shown in
Another challenge for power converters, including battery chargers, is to achieve an integrated magnetic design having light weight, low cost, and good parasitic control. Conventionally, at low frequency operation, transformer windings are usually made of copper or Litz wire. With various winding structures and core shapes, most of the wire based transformers must be assembled by hand, which can be costly and time consuming. Additionally, it can be relatively difficult to achieve good parasitic control with handmade transformers.
With the emergence of wide-band gap semiconductor devices, the switching frequency of converters can be increased by tens of times, providing the opportunity to reduce the number of turns in transformers and even the use of printed circuit board (PCB) windings. Transformers with PCB windings are also called “planar transformers” because they have a relatively low profile.
With proximity and skin effect in high frequency operation, AC winding loss can become a concern in PCB winding transformers, and a significant amount of research has been done to minimize that effect. An effective way to do so is by winding interleave, as shown in
The use of interleaved windings can result in low leakage inductance. Low leakage inductance is not a problem for certain types of power converters, such as where leakage inductance is not needed or relied upon. In certain applications, such as for LLC resonant converters and dual active bridge converters, however, an individual inductor may be required to supplement low leakage inductance. To reduce the number of magnetic components, it can be preferable to rely on transformer leakage inductance in place of such a separate inductor for low leakage inductance. To do so, the transformer leakage inductance should be a controlled and adjustable leakage inductance.
Some approaches have been proposed to increase the leakage inductance for Litz wire transformers. One approach is to change the distance between the primary winding and secondary windings to get larger leakage inductance. However, only a limited amount of increased leakage can be achieved using this method. Also, the leakage flux flows in the air, which can generate a significant amount of eddy current loss in the windings and interfere with other components. Another approach employs magnetic shunts to achieve leakage inductance. This approach does not rely on interleaving, and large AC winding loss can be expected. Also, the core loss of low permeability magnetic shunts is much larger than high permeability ones, resulting in large core loss. Additionally, magnetic shunts do not offer much ability to adjust the leakage inductance. Instead, the leakage is relatively hard to control and subject to change for large-scale production.
Other approaches rely upon the use of magnetic shunts in PCB winding transformers to create leakage inductance. The most common approach is to add a magnetic shunt layer between the primary and secondary winding layers. For example, between the primary and secondary windings, a low permeability magnetic shunt material can be added to serve as additional leakage flux path. By doing this, the leakage inductance can be significantly increased. However, there are two problems with this approach. First, it sacrifices the interleaved structure, which will increase the AC winding loss of the transformer. Second, it is difficult to realize this design in a PCB winding transformer.
According to aspects of the embodiments described herein, to improve the efficiency of the DC/DC stage in a power converter or battery charger, a variable DC-link voltage structure is proposed. In that context,
Both the AC/DC and DC/DC stages in the converter circuit 50 include an arrangement of switching transistors as shown in
In the context of efficiency,
However, due to the need for input power factor correction in the converter circuit 50, 2nd order line frequency harmonic power flows in the converter. If there is no additional control of the second DC/DC converter stage, the output current (i.e., battery charging current) can have a significant level of 2nd order line frequency ripple as shown in
In both charging and discharging grid tied modes, four control loops are relied upon in the converter circuit 50. As shown in
The control loop 110 includes a compensator 112 and a CRM/PFC controller 114. The control loop 120 includes a compensator 122. The control loop 130 includes a Simplified Optimal Trajectory Controller (SOTC) 132, a compensator 134, a voltage controlled oscillator 136, and a driver 138. The control loop 140 includes a charge profile controller 142 and a compensator 144.
Among other components, the control loops 110 120, 130, and 140 can include one or more proportional—integral (PI) and/or proportional-resonant (PR) controllers. The control loops 110, 120, 130, and 140 are configured to continuously calculate certain error values as differences between desired operating characteristics of the converter circuit 50 and measured operating characteristics (e.g., voltages, currents, frequencies, etc.) in the converter circuit 50. Based on the error values, the outputs of the control loops can be relied upon to generate switching control signals for the switching transistors in the first AC/DC and second DC/DC converter stages of the converter circuit 50. The control loops can be embodied in the form of hardware, firmware, software executable by hardware, or any combination thereof.
The control loop 110 is configured to perform CRM/PFC control for the converter circuit 50 based on an output of the control loop 120. The output of the CRM/PFC controller 114 of the control loop 110 is used to drive the switching transistors of the AC/DC stage of the converter circuit 50. The control loop 120 is configured to regulate the DC-link voltage between the AC/DC stage and the DC/DC stage in the converter circuit 50. To do so, the control loop 120 senses the voltage of the DC-link and the battery 102, and those voltages are used as reference inputs to the compensator 122 to regulate the input power. An output of the compensator 122 of the control loop 110 is provided as an input to the compensator 112 of the the control loop 120.
The compensator 134 in the control loop 130 is configured to sense the charging current, Ibat, of the battery 102 and develop a control signal to regulate the gain of the DC/DC stage through the VCO 136 by changing the frequency (e.g., for a resonant converter), phase shift angle (e.g., for a dual active bridge or phase shift full bridge), or duty cycle (e.g., for a PWM converter) of the switching control signals for the switching transistors in the DC/DC stage of the converter circuit 50. The SOTC 132 of the control loop 130 is configured to control certain transient, startup, and burst conditions for the converter circuit 50 based on the current reference Ibat. The SOTC 132 can be embodied as a controller similar to that described in U.S. Patent Publication No. 2016/0294297, the entire contents of which are hereby incorporated herein by reference. The driver 138 is configured to provide drive signals to drive the switching transistors of the DC/DC stage of the converter circuit 50.
The charge profile controller 142 of the control loop 140 is configured track the charging profile of the battery 102. The charging profile can include four stages, including a pre-charge mode, constant current charge mode, constant power charge mode, and constant voltage charge mode. In the former three modes, a current reference bat ref is provided by the charge profile controller 142 to the compensator 134. In the constant voltage charge mode, a separate voltage reference Vbat_ref can be provided as a reference to the compensator 134.
As compared to the configuration shown in
For the AC/DC stage shown in
However, neither constant on-time control nor variable on-time control may be suitable for a CRM inverter. Some control systems attempt to use hysteresis control for a CRM inverter. However, the current-sensing method in is not applicable for a multi-phase interleaved structure. Further, high-frequency instantaneous current sensing can become a bottleneck for the CRM inverter using hysteresis control. For example, the DCR sensing-derived inductor current-sensing method is not applicable, since the common-mode voltage across the inductor is too large in offline applications and a sensing resistor in series with the bottom switch does not work in the negative line cycle. Thus, the sensing resistor in the return path is suitable for single-phase topology but not a multiphase topology with interleaving. Further, the current transformer (CT) method can be applied applicable, but each high-frequency switch needs one CT connected in series, which makes the critical power loop very large. That can induce significant switching loss and parasitic ringing.
In order to avoid instantaneous current sensing, a hybrid control strategy combining average current control and part hysteresis control is proposed, as illustrated in
The zero crossing detector 206, programmed timer 208, and comparator 210 are configured to control the switching cycle CRM operation to achieve zero voltage switching (ZVS) through the entire line cycle. The zero crossing detector 206, programmed timer 208, and comparator 210 are thus configured to control the minimum current in the inductor L, shown as IL_neg in
The proportional unit 204 and the compensator 112 are configured to control the average current in the inductor L to track the desired sinusoidal reference, shown as IL_avg in
Another benefit of the variable DC-link voltage concepts described herein is that the resonant inductor can be relatively smaller as compared as to the conventional use of fixed DC-link voltages. This provides the opportunity to use PCB winding transformers and the leakage inductance of such transformers as a resonant inductor.
To achieve the desired leakage inductance with a PCB winding transformer without sacrificing a significant amount of AC winding loss, a new E-I transformer structure is proposed. In that context,
The transformer 300 is different than conventional E-I transformers. Windings are placed on the outer posts 311 and 312 of the core 310 instead of the center post 313. Further, the volume or area of the center post 313 is not two times that of either of the outer posts 311 or 312, but can be adjusted according to the desired leakage inductance. Another difference is that the windings are re-distributed as compared to conventional E-I transformers. In other words, on the two outer posts 311 and 312, the number of the primary windings 320 and the secondary windings 322 are not distributed equally.
Additionally, the center post 313 is used as a leakage path. By doing so, leakage flux is created without much change in the interleaving between the primary winding 320 and secondary winding 322. With this feature, the leakage inductance of the transformer 300 can be controlled by adjusting the reluctance of the center post 313. By increasing the reluctance of the center post 313, less leakage flux will flow through center post 313. Thus, the leakage of the transformer 300 can be reduced. By reducing the reluctance of the center post 313, more leakage flux will flow through center post 313. Thus, the leakage of the transformer 300 can be increased.
If the T-type transformer equivalent circuit shown in
where Rg1 and Rg2 are the reluctance of the outer post 311 and 312 air gap and the center post 313 air gap, respectively.
Equations (2) and (3) mathematically show that, by changing the outer post 311 and 312 and center post 313 air gap reluctance, the magnetizing inductance and leakage inductance of the transformer 300 can be adjusted individually. In practice, if the leakage inductance of the transformer 300 needs to be increased, then the center post air gap can be increased by reducing the length of the center post 313. If the leakage inductance of the transformer 300 needs to be reduced, then the center post air gap can be reduced by increasing the length of the center post air 313.
The proposed concept can be extended to a more generalized concept, as shown in
With reference to
From the above equations, it can be seen that the more unbalanced the winding distribution (e.g., the less interleaving of the windings), the more leakage by changing the air gap reluctance. So, for different applications, different winding distributions and different air gaps can be selected to meet the required leakage inductance.
The concept can also be extended to a transformer having more than two windings. In that context,
With this transformer equation, the leakage inductance and the magnetizing inductance can be determined.
The components described herein, including the control loops 110, 120, 130, and 140 can be embodied in the form of hardware, firmware, software executable by hardware, or as any combination thereof. If embodied as hardware, the components described herein can be implemented as a collection of discrete analog, digital, or mixed analog and digital circuit components. The hardware can include one or more discrete logic circuits, microprocessors, microcontrollers, or digital signal processors (DSPs), application specific integrated circuits (ASICs), programmable logic devices (e.g., field-programmable gate array (FPGAs)), or complex programmable logic devices (CPLDs)), among other types of processing circuitry.
The microprocessors, microcontrollers, or DSPs, for example, can execute software to perform the control aspects of the embodiments described herein. Any software or program instructions can be embodied in or on any suitable type of non-transitory computer-readable medium for execution. Example computer-readable mediums include any suitable physical (i.e., non-transitory or non-signal) volatile and non-volatile, random and sequential access, read/write and read-only, media, such as hard disk, floppy disk, optical disk, magnetic, semiconductor (e.g., flash, magneto-resistive, etc.), and other memory devices. Further, any component described herein can be implemented and structured in a variety of ways. For example, one or more components can be implemented as a combination of discrete and integrated analog and digital components.
The above-described examples of the present disclosure are merely possible examples of implementations set forth for a clear understanding of the principles of the disclosure. Many variations and modifications can be made without departing substantially from the spirit and principles of the disclosure. All such modifications and variations are intended to be included herein within the scope of this disclosure and protected by the following claims.