SIGNAL GENERATION METHOD, TRANSMISSION DEVICE, RECEPTION METHOD, AND RECEPTION DEVICE

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First Claim
1. A transmission method used in a transmission system that includes a first transmission station and a second transmission station, the transmission method comprising:
 performing, by the first transmission station, first phase changing on signals included in a first orthogonal frequencydivision multiplexing (OFDM) frame according to a first phase changing pattern or a second phase changing pattern;
performing, by the second transmission station, second phase changing on signals included in a second OFDM frame according to a third phase changing pattern or a fourth phase changing pattern, the second OFDM frame being identical to the first OFDM frame;
converting, by the first transmission station, a first control information modulated signals to generate a first preamble, and converting, by the first transmission station, the first OFDM frame to generate a first OFDM signal, the first control information modulated signals being generated from control information;
converting, by the second transmission station, a second control information modulated signals to generate a second preamble, and converting, by the second transmission station, the second OFDM frame to generate a second OFDM signal, the second control information modulated signals being identical to the first control information modulated signals;
transmitting, by the first transmission station, the first preamble and the first OFDM signal; and
transmitting, by the second transmission station, the second preamble and the second OFDM signal, whereinthe control information includes information indicating the phase changing patterns used for the first phase changing and the second phase changing, andthe first preamble is generated without undergoing the first phase changing, and the second preamble is generated without undergoing the second phase changing, andthe first OFDM frame includes modulated signals generated by using a modulation scheme having N×
N candidate signal points, a real component value of each candidate signal point is one from among N candidate values, an imaginary component value of each candidate signal point is one from among the N candidate values, wherein N is a positive integer greater than three that is also a power of two, andthe N candidate values include at least a first value, a second value which is lower than and next to the first value, and a third value which is higher than and next to the first value, a distance between the first value and the second value is different from a distance between the first value and the third value, andN is 64.
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Abstract
A signal generation method is used in a transmission device that transmits a plurality of transmission signals from a plurality of antennas at the same frequency and at the same time, in the case where larger power change is performed on a first transmission signal than on a second transmission signal during generation process of the first transmission signal and the second transmission signal, the first transmission signal and the second transmission signal are mapped before the power change such that a minimum Euclidian distance between possible signal points for the first signal is longer than a minimum Euclidian distance between possible signal points for the second signal.
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4 Claims
 1. A transmission method used in a transmission system that includes a first transmission station and a second transmission station, the transmission method comprising:
performing, by the first transmission station, first phase changing on signals included in a first orthogonal frequencydivision multiplexing (OFDM) frame according to a first phase changing pattern or a second phase changing pattern; performing, by the second transmission station, second phase changing on signals included in a second OFDM frame according to a third phase changing pattern or a fourth phase changing pattern, the second OFDM frame being identical to the first OFDM frame; converting, by the first transmission station, a first control information modulated signals to generate a first preamble, and converting, by the first transmission station, the first OFDM frame to generate a first OFDM signal, the first control information modulated signals being generated from control information; converting, by the second transmission station, a second control information modulated signals to generate a second preamble, and converting, by the second transmission station, the second OFDM frame to generate a second OFDM signal, the second control information modulated signals being identical to the first control information modulated signals; transmitting, by the first transmission station, the first preamble and the first OFDM signal; and transmitting, by the second transmission station, the second preamble and the second OFDM signal, wherein the control information includes information indicating the phase changing patterns used for the first phase changing and the second phase changing, and the first preamble is generated without undergoing the first phase changing, and the second preamble is generated without undergoing the second phase changing, and the first OFDM frame includes modulated signals generated by using a modulation scheme having N×
N candidate signal points, a real component value of each candidate signal point is one from among N candidate values, an imaginary component value of each candidate signal point is one from among the N candidate values, wherein N is a positive integer greater than three that is also a power of two, andthe N candidate values include at least a first value, a second value which is lower than and next to the first value, and a third value which is higher than and next to the first value, a distance between the first value and the second value is different from a distance between the first value and the third value, and N is 64.
 2. A transmission system that includes a first transmission station and a second transmission station, wherein
the first transmission station comprises: a first phase changer that, in operation, performs first phase changing on signals included in a first orthogonal frequencydivision multiplexing (OFDM) frame according to a first phase changing pattern or a second phase changing pattern; a first inverse fast fourier transform (IFFT) unit that, in operation, converts a first control information modulated signals to generate a first preamble, and converts the first OFDM frame to generate a first OFDM signal, the first control information modulated signals being generated from control information; and a first antenna that, in operation, transmits the first preamble and the first OFDM signal; the second transmission station comprises; a second phase changer that, in operation, performs second phase changing on signals included in a second OFDM frame according to a third phase changing pattern or a fourth phase changing pattern, the second OFDM frame being identical to the first OFDM frame; a second IFFT unit that, in operation, converts a second control information modulated signals to generate a second preamble, and converts the second OFDM frame to generate a second OFDM signal, the second control information modulated signals being identical to the first control information modulated signals; and a first antenna that, in operation, transmits the second preamble and the second OFDM signal, wherein the control information includes information indicating the phase changing patterns used for the first phase changing and the second phase changing, and the first preamble is generated without undergoing the first phase changing, and the second preamble is generated without undergoing the second phase changing, and the first OFDM frame includes modulated signals generated by using a modulation scheme having N×
N candidate signal points, a real component value of each candidate signal point is one from among N candidate values, an imaginary component value of each candidate signal point is one from among the N candidate values, wherein N is a positive integer greater than three that is also a power of two, andthe N candidate values include at least a first value, a second value which is lower than and next to the first value, and a third value which is higher than and next to the first value, a distance between the first value and the second value is different from a distance between the first value and the third value, and N is 64.
 3. A reception method used in a reception device that receives a signal transmitted from a transmission system, the reception method comprising:
receiving a first reception signal obtained by receiving a first preamble and a second preamble transmitted from a first antenna and a second antenna respectively, and receiving a second reception signal obtained by receiving a first orthogonal frequencydivision multiplexing (OFDM) signal and a second OFDM signal transmitted from the first antenna and the second antenna respectively, wherein the first preamble is generated by converting a first control information modulated signals into the first preamble, the first control information modulated signals being generated from control information, and the second preamble is generated by converting a second control information modulated signals into the second preamble, the second control information modulated signals are identical to the first control information modulated signals, and the first OFDM signal is generated by performing first phase changing on signals included in a first OFDM frame according to a first phase changing pattern or a second phase changing pattern, converting the first OFDM frame into the first OFDM signal, and the second OFDM signal is generated by performing first phase changing on signals included in a first OFDM frame according to a third phase changing pattern or a fourth phase changing pattern, converting the second OFDM frame into the second OFDM signal, the second OFDM frame being identical to the first OFDM frame; and demodulating the second reception signal based on the control information acquired from the first reception signal, wherein the control information includes information indicating the phase changing patterns used for the first phase changing and the second phase changing, and the first preamble is generated without undergoing the first phase changing, and the second preamble is generated without undergoing the second phase changing, and the first OFDM frame includes modulated signals generated by using a modulation scheme having N×
N candidate signal points, a real component value of each candidate signal point is one from among N candidate values, an imaginary component value of each candidate signal point is one from among the N candidate values, wherein N is a positive integer greater than three that is also a power of two, andthe N candidate values include at least a first value, a second value which is lower next to the first value, and a third value which is higher than and next to the first value, a distance between the first value and the second value is different from a distance between the first value and the third value, and N is 64.
 4. A reception device that receives a signal transmitted from a transmission system, the reception device comprising:
a receiver that, in operation, receives a first reception signal and a second reception signal, the first reception signal being a signal obtained by receiving a first preamble and a second preamble transmitted from a first antenna and a second antenna respectively, the second reception signal being a signal obtained by receiving a first orthogonal frequencydivision multiplexing (OFDM) signal and a second OFDM signal transmitted from the first antenna and the second antenna respectively, wherein the first preamble is generated by converting a first control information signals into the first preamble, the first control information modulated signals being generated from control information, and the second preamble is generated by converting a second control information modulated signals into the second preamble, the second control information modulated signals are identical to the first control information modulated signals, and the first OFDM signal is generated by performing first phase changing on signals included in a first OFDM frame according to a first phase changing pattern or a second phase changing pattern, converting the first OFDM frame into the first OFDM signal, and the second OFDM signal is generated by performing first phase changing on signals included in a first OFDM frame according to a third phase changing pattern or a fourth phase changing pattern, converting the second OFDM frame into the second OFDM signal, the second OFDM frame being identical to the first OFDM frame; and a demodulator that, in operation, demodulates the second reception signal based on the control information acquired from the first reception signal, wherein the control information includes information indicating the phase changing patterns used for the first phase changing and the second phase changing, and the first preamble is generated without undergoing the first phase changing, and the second preamble is generated without undergoing the second phase changing, and the first OFDM frame includes modulated signals generated by using a modulation scheme having N×
N candidate signal points, a real component value of each candidate signal point is one from among N candidate values, an imaginary component value of each candidate signal point is one from among the N candidate values, wherein N is a positive integer greater than three that is also a power of two, andthe N candidate values include at least a first value, a second value which is lower next to the first value, and a third value which is higher than and next to the first value, a distance between the first value and the second value is different from a distance between the first value and the third value, and N is 64.
1 Specification
This application is based on the application No. 2012268858 filed Dec. 7, 2012 and the application No. 2012268859 filed Dec. 7, 2012 in Japan, the claims, the specification, the drawings, and the abstract of which are hereby incorporated by reference.
The present invention relates to a transmission device and a reception device for communication using multiple antennas.
A MIMO (MultipleInput, MultipleOutput) system is an example of a conventional communication system using multiple antennas. In multiantenna communication, of which the MIMO system is typical, multiple transmission signals are each modulated, and each modulated signal is simultaneously transmitted from a different antenna in order to increase the transmission speed of the data.
In this context, Patent Literature 1 suggests using a transmission device provided with a different interleaving pattern for each transmit antenna. That is, the transmission device from
As it happens, models of actual propagation environments in wireless communications include NLOS (Non LineOfSight), typified by a Rayleigh fading environment is representative, and LOS (LineOfSight), typified by a Rician fading environment. When the transmission device transmits a single modulated signal, and the reception device performs maximal ratio combination on the signals received by a plurality of antennas and then demodulates and decodes the resulting signals, excellent reception quality can be achieved in a LOS environment, in particular in an environment where the Rician factor is large. The Rician factor represents the received power of direct waves relative to the received power of scattered waves. However, depending on the transmission system (e.g., a spatial multiplexing MIMO system), a problem occurs in that the reception quality deteriorates as the Rician factor increases (see NonPatent Literature 3).
Broadcast or multicast communication is a service applied to various propagation environments. The radio wave propagation environment between the broadcaster and the receivers belonging to the users is often a LOS environment. When using a spatial multiplexing MIMO system having the above problem for broadcast or multicast communication, a situation may occur in which the received electric field strength is high at the reception device, but in which degradation in reception quality makes service reception difficult. In other words, in order to use a spatial multiplexing MIMO system in broadcast or multicast communication in both the NLOS environment and the LOS environment, a MIMO system that offers a certain degree of reception quality is desirable.
NonPatent Literature 8 describes a scheme for selecting a codebook used in precoding (i.e. a precoding matrix, also referred to as a precoding weight matrix) based on feedback information from a communication party. However, NonPatent Literature 8 does not at all disclose a scheme for precoding in an environment in which feedback information cannot be acquired from the other party, such as in the above broadcast or multicast communication.
On the other hand, NonPatent Literature 4 discloses a scheme for switching the precoding matrix over time. This scheme is applicable when no feedback information is available. NonPatent Literature 4 discloses using a unitary matrix as the precoding matrix, and switching the unitary matrix at random, but does not at all disclose a scheme applicable to degradation of reception quality in the abovedescribed LOS environment. NonPatent Literature 4 simply recites hopping between precoding matrices at random. Obviously, NonPatent Literature 4 makes no mention whatsoever of a precoding method, or a structure of a precoding matrix, for remedying degradation of reception quality in a LOS environment.
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An object of the present invention is to provide a MIMO system that improves reception quality in a LOS environment.
The present invention provides a signal generation method for use in a transmission device that transmits a plurality of transmission signals from a plurality of antennas at the same frequency and at the same time, the signal generation method comprising: generating a first modulated signal s_{1}(i) from first transmission data of g bits, and generating a second modulated signal s_{2}(i) from second transmission data of h bits; and generating a first signal z_{1}(i) and a second signal z_{2}(i) that satisfy the following formula R2 from the first modulated signal s_{1}(i) and the second modulated signal s_{2}(i), where a(i), b(i), c(i), and d(i) each denote an arbitrary complex number, at least two of a(i), b(i), c(i), and d(i) each denote a value other than zero, P_{1 }and P_{2 }each denote a real number, and Q_{1 }and Q_{2 }each denote a real number and satisfy Q_{1}>Q_{2}, and when a third signal u_{1}(i) and a fourth signal u_{2}(i) are defined such that z_{1}(i)=Q_{1}×u_{1}(i) and z_{2}(i)=Q_{2}×u_{2}(i) are satisfied, D_{1}>D_{2 }is satisfied, where D_{1 }represents a minimum Euclidian distance between 2^{g+h }possible signal points for the third signal u_{1}(i) in an I (inphase)Q (quadrature) plane, and D_{2 }represents a minimum Euclidian distance between 2^{g+h }possible signal points for the fourth signal u_{2}(i) in an I (inphase)Q (quadrature) plane.
Also, the present invention provides a transmission device that transmits a plurality of transmission signals from a plurality of antennas at the same frequency and at the same time, the transmission device comprising: a mapper generating a first modulated signal s_{1}(i) from first transmission data of g bits, and generating a second modulated signal s_{2}(i) from second transmission data of h bits; and a weighting unit generating a first signal z_{1}(i) and a second signal z_{2}(i) that satisfy the following formula R2 from the first modulated signal s_{1}(i) and the second modulated signal s_{2}(i), where a(i), b(i), c(i), and d(i) each denote an arbitrary complex number, at least two of a(i), b(i), c(i), and d(i) each denote a value other than zero, P_{1 }and P_{2 }each denote a real number, and Q_{1 }and Q_{2 }each denote a real number and satisfy Q_{1}>Q_{2}, and when a third signal u_{1}(i) and a fourth signal u_{2}(i) are defined such that z_{1}(i)=Q_{1}×u_{1}(i) and z_{2}(i)=Q_{2}×u_{2}(i) are satisfied, D_{1}>D_{2 }is satisfied, where D_{1 }represents a minimum Euclidian distance between 2^{g+h }possible signal points for the third signal u_{1}(i) in an I (inphase)Q (quadrature) plane, and D_{2 }represents a minimum Euclidian distance between 2^{g+h }possible signal points for the fourth signal u_{2}(i) in an I (inphase)Q (quadrature) plane.
According to the above structure, the present invention provides a signal generation method and a signal generation apparatus that remedy degradation of reception quality in a LOS environment, thereby providing highquality service to LOS users during broadcast or multicast communication.
Embodiments of the present invention are described below with reference to the accompanying drawings.
The following describes, in detail, a transmission scheme, a transmission device, a reception scheme, and a reception device pertaining to the present embodiment.
Before beginning the description proper, an outline of transmission schemes and decoding schemes in a conventional spatial multiplexing MIMO system is provided.
Here, H_{NtNr }is the channel matrix, n=(n_{1}, . . . , n_{Nr}) is the noise vector, and the average value of n_{i }is zero for independent and identically distributed (i.i.d) complex Gaussian noise of variance σ^{2}. Based on the relationship between transmitted symbols introduced into a receiver and the received symbols, the probability distribution of the received vectors can be expressed as formula 2, below, for a multidimensional Gaussian distribution.
Here, a receiver performing iterative decoding is considered. Such a receiver is illustrated in
The following describes the MIMO signal iterative detection performed by the N_{t}×N_{r }spatial multiplexing MIMO system.
The loglikelihood ratio of u_{mn }is defined by formula 6.
Through application of Bayes'"'"' theorem, formula 6 can be expressed as formula 7.
Note that U_{mn,±1}={(uu_{mn}=±+1}. Through the approximation ln Σa_{j}˜ max In a_{j}, formula 7 can be approximated as formula 8. The symbol ˜ is herein used to signify approximation.
In formula 8, P(uu_{mn}) and In P(uu_{mn}) can be expressed as follows.
Note that the logprobability of the formula given in formula 2 can be expressed as formula 12.
Accordingly, given formula 7 and formula 13, the posterior Lvalue for the MAP or APP (a posteriori probability) can be can be expressed as follows.
This is hereinafter termed iterative APP decoding. Also, given formula 8 and formula 12, the posterior Lvalue for the Maxlog APP can be can be expressed as follows.
This is hereinafter referred to as iterative Maxlog APP decoding. As such, the external information required by the iterative decoding system is obtainable by subtracting prior input from formula 13 or from formula 14.
The receiver performs iterative detection (iterative APP (or Maxlog APP) decoding) of MIMO signals, as described above. The LDPC codes are decoded using, for example, sumproduct decoding.
[Math. 16]
(i_{a},j_{a})=π_{a}(Ω_{ia,ja}^{a}) (formula 16)
[Math. 17]
(i_{b},j_{b})=π_{b}(Ω_{ib,jb}^{a}) (formula 17)
Here, i_{a }and i_{b }represent the symbol order after interleaving, j_{a }and j_{b }represent the bit position in the modulation scheme (where j_{a},j_{b}=1, . . . , h), π_{a }and π_{b }represent the interleavers of streams A and B, and Ω^{a}_{ia,ja }and Q^{b}_{ib,jb }represent the data order of streams A and B before interleaving. Note that
The following describes, in detail, the sumproduct decoding used in decoding the LDPC codes and the MIMO signal iterative detection algorithm, both used by the receiver.
SumProduct Decoding
A twodimensional M×N matrix H={H_{mn}} is used as the check matrix for LDPC codes subject to decoding. For the set[1,N]={1, 2, . . . , N}, the partial sets A(m) and B(n) are defined as follows.
[Math. 18]
A(m)≡{n:H_{mn}=1} (formula 18)
[Math. 19]
B(n)≡{m:H_{mn}=1} (formula 19)
Here, A(m) signifies the set of column indices equal to 1 for row m of check matrix H, while B(n) signifies the set of row indices equal to 1 for row n of check matrix H. The sumproduct decoding algorithm is as follows.
Step A1 (Initialization): For all pairs (m,n) satisfying H_{mn}=1, set the prior log ratio β_{mn}=1. Set the loop variable (number of iterations) l_{sum}=1, and set the maximum number of loops l_{sum,max}.
Step A2 (Processing): For all pairs (m,n) satisfying H_{mn}=1 in the order m=1, 2, . . . , M, update the extrinsic value log ratio Umn using the following update formula.
where f is the Gallager function. λ_{n }can then be computed as follows.
Step A3 (Column Operations): For all pairs (m,n) satisfying H_{mn}=1 in the order n=1, 2, . . . , N, update the extrinsic value log ratio β_{mn }using the following update formula.
Step A4 (Loglikelihood Ratio Calculation): For n∈[1,N], the loglikelihood ratio L_{n }is computed as follows.
Step A5 (Iteration Count): If l_{sum}<l_{sum,max}, then l_{sum }is incremented and the process returns to step A2. Sumproduct decoding ends when l_{sum}=l_{sum,max}.
The above describes one iteration of sumproduct decoding operations. Afterward, MIMO signal iterative detection is performed. The variables m, n, α_{mn}, β_{mn}, λ_{n}, and L_{n }used in the above explanation of sumproduct decoding operations are expressed as m_{a}, n_{a}, α^{a}_{mana}, β^{a}_{mana}, λ_{na}, and L_{na }for stream A and as m_{b}, n_{b}, α^{b}_{mbnb}, β^{b}_{mbnb}, λ_{nb}, and L_{nb }for stream B.
The following describes the calculation of λ_{n }for MIMO signal iterative detection.
The following formula is derivable from formula 1.
Given the frame configuration illustrated in
[Math. 26]
n_{a}=Ω_{ia,ja}^{a} (formula 26)
[Math. 27]
n_{b}=Ω_{ib,jb}^{b} (formula 27)
where n_{a},n_{b }∈[1,N]. For iteration k of MIMO signal iterative detection, the variables λ_{na}, L_{na}, λ_{nb}, and L_{nb }are expressed as λ_{k,na}, L_{k,na}, λ_{κ,nb}, and L_{k,nb}.
Step B1 (Initial Detection; k=0): For initial wave detection, λ_{o,na }and λ_{0,nb }are calculated as follows.
For iterative APP decoding:
For iterative Maxlog APP decoding:
where X=a,b. Next, the iteration count for the MIMO signal iterative detection is set to l_{mimo}=0, with the maximum iteration count being l_{mimo,max}.
Step B2 (Iterative Detection; Iteration k): When the iteration count is k, formula 11, formula 13) through formula 15), formula 16), and formula 17) can be expressed as formula 31) through formula 34), below. Note that (X,Y)=(a,b)(b,a).
For iterative APP decoding:
For iterative Maxlog APP decoding:
Step B3 (Iteration Count and Codeword Estimation): If l_{mimo}<l_{mimo,max}, then l_{mimo }is incremented and the process returns to step B2. When l_{mimo}=l_{mimo,max}, an estimated codeword is found, as follows.
where X=a,b.
An interleaver 304A takes the encoded data 303A and the frame configuration signal 313 as input, performs interleaving, i.e., rearranges the order thereof, and then outputs interleaved data 305A. (Depending on the frame configuration signal 313, the interleaving scheme may be switched.)
A mapper 306A takes the interleaved data 305A and the frame configuration signal 313 as input and performs modulation, such as QPSK (Quadrature Phase Shift Keying), 16QAM (16Quadradature Amplitude Modulation), or 64QAM (64Quadradture Amplitude Modulation) thereon, then outputs a baseband signal 307A. (Depending on the frame configuration signal 313, the modulation scheme may be switched.)
An encoder 302B takes information (data) 301B and the frame configuration signal 313 as input (which includes the errorcorrection scheme, coding rate, block length, and other information used by the encoder 302A in errorcorrection coding of the data, such that the scheme designated by the frame configuration signal 313 is used. The errorcorrection scheme may be switched). In accordance with the frame configuration signal 313, the encoder 302B performs errorcorrection coding, such as convolutional encoding, LDPC encoding, turbo encoding or similar, and outputs encoded data 303B.
An interleaver 304B takes the encoded data 303B and the frame configuration signal 313 as input, performs interleaving, i.e., rearranges the order thereof, and outputs interleaved data 305B. (Depending on the frame configuration signal 313, the interleaving scheme may be switched.)
A mapper 306B takes the interleaved data 305B and the frame configuration signal 313 as input and performs modulation, such as QPSK, 16QAM, or 64QAM thereon, then outputs a baseband signal 307B. (Depending on the frame configuration signal 313, the modulation scheme may be switched.)
A signal processing scheme information generator 314 takes the frame configuration signal 313 as input and accordingly outputs signal processing scheme information 315. The signal processing scheme information 315 designates the fixed precoding matrix to be used, and includes information on the pattern of phase changes used for changing the phase.
A weighting unit 308A takes baseband signal 307A, baseband signal 307B, and the signal processing scheme information 315 as input and, in accordance with the signal processing scheme information 315, performs weighting on the baseband signals 307A and 307B, then outputs a weighted signal 309A. The weighting scheme is described in detail, later.
A wireless unit 310A takes weighted signal 309A as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on, then outputs transmit signal 311A. Transmit signal 311A is then output as radio waves by an antenna 312A.
A weighting unit 308B takes baseband signal 307A, baseband signal 307B, and the signal processing scheme information 315 as input and, in accordance with the signal processing scheme information 315, performs weighting on the baseband signals 307A and 307B, then outputs weighted signal 316B.
Both weighting units perform weighting using a fixed precoding matrix. The precoding matrix uses, for example, the scheme of formula 36, and satisfies the conditions of formula 37 or formula 38, all found below. However, this is only an example. The value of α is not restricted to formula 37 and formula 38, and may take on other values, e.g., α=1.
Here, the precoding matrix is:
In formula 36,
α may be given by formula 37.
Alternatively, in formula 36,
α may be given by formula 38.
The precoding matrix is not restricted to that of formula 36, but may also be as indicated by formula 39.
In formula 39, let a=Ae^{jδ11}, b=Be^{jδ12}, c=Ce^{jδ21}, and d=De^{jδ22}. Further, one of a, b, c, and d may be zero. For example, the following configurations are possible: (1) a may be zero while b, c, and d are nonzero, (2) b may be zero while a, c, and d are nonzero, (3) c may be zero while a, b, and d are nonzero, or (4) d may be zero while a, b, and c are nonzero.
When any of the modulation scheme, errorcorrecting codes, and the coding rate thereof are changed, the precoding matrix may also be set, changed, and fixed for use.
A phase changer 317B takes weighted signal 316B and the signal processing scheme information 315 as input, then regularly changes the phase of the signal 316B for output. This regular change is a change of phase performed according to a predetermined phase changing pattern having a predetermined period (cycle) (e.g., every n symbols (n being an integer, n>1) or at a predetermined interval). The details of the phase changing pattern are explained below, in Embodiment 4.
Wireless unit 310B takes postphasechange signal 309B as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on, then outputs transmit signal 311B. Transmit signal 311B is then output as radio waves by an antenna 312B.
An encoder 402 takes information (data) 401 and the frame configuration signal 313 as input, and, in accordance with the frame configuration signal 313, performs errorcorrection coding and outputs encoded data 402.
A distributor 404 takes the encoded data 403 as input, performs distribution thereof, and outputs data 405A and data 405B. Although
Symbol 501_1 is for estimating channel fluctuations for modulated signal z1(t) (where t is time) transmitted by the transmission device. Symbol 502_1 is a data symbol transmitted by modulated signal z1(t) as symbol number u (in the time domain). Symbol 503_1 is a data symbol transmitted by modulated signal z1(t) as symbol number u+1.
Symbol 5012 is for estimating channel fluctuations for modulated signal z2(t) (where t is time) transmitted by the transmission device. Symbol 502_2 is a data symbol transmitted by modulated signal z2(t) as symbol number u (in the time domain). Symbol 503_2 is a data symbol transmitted by modulated signal z1(t) as symbol number u+1.
Here, the symbols of z1(t) and of z2(t) having the same time (identical timing) are transmitted from the transmit antenna using the same (shared/common) frequency.
The following describes the relationships between the modulated signals z1(t) and z2(t) transmitted by the transmission device and the received signals r1(t) and r2(t) received by the reception device.
In
Here, given vector W1=(w11,w12) from the first row of the fixed precoding matrix F, z1(t) is expressible as formula 41, below.
[Math. 41]
z1(t)=W1×(s1(t),s2(t))^{T} (formula 41)
Similarly, given vector W2=(w21,w22) from the second row of the fixed precoding matrix F, and letting the phase changing formula applied by the phase changer by y(t), then z2(t) is expressible as formula 42, below.
[Math. 42]
z2(t)=y(t)×W2×(s1(t),s2(t))^{T} (formula 42)
Here, y(t) is a phase changing formula following a predetermined scheme. For example, given a period (cycle) of four and time u, the phase changing formula is expressible as formula 43, below.
[Math. 43]
y(u)=e^{j0} (formula 43)
Similarly, the phase changing formula for time u+1 may be, for example, as given by formula 44.
That is, the phase changing formula for time u+k is expressible as formula 45.
Note that formula 43 through formula 45 are given only as an example of regular phase changing.
The regular change of phase is not restricted to a period (cycle) of four. Improved reception capabilities (the errorcorrection capabilities, to be exact) may potentially be promoted in the reception device by increasing the period (cycle) number (this does not mean that a greater period (cycle) is better, though avoiding small numbers such as two is likely ideal).
Furthermore, although formula 43 through formula 45, above, represent a configuration in which a change in phase is carried out through rotation by consecutive predetermined phases (in the above formula, every 7/2), the change in phase need not be rotation by a constant amount, but may also be random. For example, in accordance with the predetermined period (cycle) of y(t), the phase may be changed through sequential multiplication as shown in formula 46 and formula 47. The key point of regular phase changing is that the phase of the modulated signal is regularly changed. The degree of phase change is preferably as even as possible, such as from −π radians to π radians. However, given that this describes a distribution, random changes are also possible.
As such, the weighting unit 600 of
When a specialized precoding matrix is used in a LOS environment, the reception quality is likely to improve tremendously. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may greatly differ from the specialized precoding matrix, upon reception. The LOS environment has certain rules. Thus, data reception quality is tremendously improved through a regular change applied to a transmit signal that obeys those rules. The present invention offers a signal processing scheme for improvements in the LOS environment.
Channel fluctuation estimator 705_1 for modulated signal z1 transmitted by the transmission device takes baseband signal 704_X as input, extracts reference symbol 501_1 for channel estimation from
Channel fluctuation estimator 705_2 for modulated signal z2 transmitted by the transmission device takes baseband signal 704_X as input, extracts reference symbol 501_2 for channel estimation from
Wireless unit 703_Y receives, as input, received signal 702Y received by antenna 701_X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal 704_Y.
Channel fluctuation estimator 707_1 for modulated signal z1 transmitted by the transmission device takes baseband signal 704_Y as input, extracts reference symbol 501_1 for channel estimation from
Channel fluctuation estimator 707_2 for modulated signal z2 transmitted by the transmission device takes baseband signal 704_Y as input, extracts reference symbol 501_2 for channel estimation from
A control information decoder 709 receives baseband signal 704_X and baseband signal 704_Y as input, detects symbol 500_1 that indicates the transmission scheme from
A signal processor 711 takes the baseband signals 704_X and 704_Y, the channel estimation signals 706_1, 706_2, 708_1, and 7082, and the transmission scheme information signal 710 as input, performs detection and decoding, and then outputs received data 712_1 and 712_2.
Next, the operations of the signal processor 711 from
Here, the reception device may use the decoding schemes of NonPatent Literature 2 and 3 on R(t) by computing H(t)×Y(t)×F.
Accordingly, the coefficient generator 819 from
The inner MIMO detector 803 takes the signal processing scheme information signal as input and performs iterative detection and decoding using the signal and the relationship thereof to formula 48. The operations thereof are described below.
The processor illustrated in
In
Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection).
(Initial Detection)
The inner MIMO detector 803 takes baseband signal 801X, channel estimation signal group 802X, baseband signal 801Y, and channel estimation signal group 802Y as input. Here, the modulation scheme for modulated signal (stream) s1 and modulated signal (stream) s2 is taken to be 16QAM.
The inner MIMO detector 803 first computes H(t)×Y(t)×F from the channel estimation signal groups 802X and 802Y, thus calculating a candidate signal point corresponding to baseband signal 801X.
Similarly, the inner MIMO detector 803 computes H(t)×Y(t)×F from the channel estimation signal groups 802X and 802Y, calculates candidate signal points corresponding to baseband signal 801Y, computes the Euclidean squared distance between each of the candidate signal points and the received signal points (corresponding to baseband signal 801Y), and divides the Euclidean squared distance by the noise variance σ^{2}. Accordingly, E_{Y}(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. That is, E_{Y }is the Euclidian squared distance between a candidate signal point corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance.
Next, E_{X}(b0, b1, b2, b3, b4, b5, b6, b7)+E_{Y}(b0, b1, b2, b3, b4, b5, b6, b7)=E(b0, b1, b2, b3, b4, b5, b6, b7) is computed.
The inner MIMO detector 803 outputs E(b0, b1, b2, b3, b4, b5, b6, b7) as a signal 804.
Loglikelihood calculator 805A takes the signal 804 as input, calculates the loglikelihood of bits b0, b1, b2, and b3, and outputs loglikelihood signal 806A. Note that this loglikelihood calculation produces the loglikelihood of a bit being 1 and the loglikelihood of a bit being 0. The calculation scheme is as shown in formula 28, formula 29, and formula 30, and the details are given by NonPatent Literature 2 and 3.
Similarly, loglikelihood calculator 805A takes the signal 804 as input, calculates the loglikelihood of bits b0, b1, b2, and b3, and outputs loglikelihood signal 806B.
A deinterleaver (807A) takes loglikelihood signal 806A as input, performs deinterleaving corresponding to that of the interleaver (the interleaver (304A) from
Similarly, a deinterleaver (807B) takes loglikelihood signal 806B as input, performs deinterleaving corresponding to that of the interleaver (the interleaver (304B) from
Loglikelihood ratio calculator 809A takes deinterleaved loglikelihood signal 808A as input, calculates the loglikelihood ratio of the bits encoded by encoder 302A from
Similarly, loglikelihood ratio calculator 809B takes deinterleaved loglikelihood signal 808B as input, calculates the loglikelihood ratio of the bits encoded by encoder 302B from
Softin/softout decoder 811A takes loglikelihood ratio signal 810A as input, performs decoding, and outputs decoded loglikelihood ratio 812A.
Similarly, softin/softout decoder 811B takes loglikelihood ratio signal 810B as input, performs decoding, and outputs decoded loglikelihood ratio 812B.
(Iterative Decoding (Iterative Detection), k Iterations)
The interleaver (813A) takes the k−1th decoded loglikelihood ratio 812A decoded by the softin/softout decoder as input, performs interleaving, and outputs interleaved loglikelihood ratio 814A. Here, the interleaving pattern used by the interleaver (813A) is identical to that of the interleaver (304A) from
Another interleaver (813B) takes the k−1th decoded loglikelihood ratio 812B decoded by the softin/softout decoder as input, performs interleaving, and outputs interleaved loglikelihood ratio 814B. Here, the interleaving pattern used by the other interleaver (813B) is identical to that of another interleaver (304B) from
The inner MIMO detector 803 takes baseband signal 816X, transformed channel estimation signal group 817X, baseband signal 816Y, transformed channel estimation signal group 817Y, interleaved loglikelihood ratio 814A, and interleaved loglikelihood ratio 814B as input. Here, baseband signal 816X, transformed channel estimation signal group 817X, baseband signal 816Y, and transformed channel estimation signal group 817Y are used instead of baseband signal 801X, channel estimation signal group 802X, baseband signal 801Y, and channel estimation signal group 802Y because the latter cause delays due to the iterative decoding.
The iterative decoding operations of the inner MIMO detector 803 differ from the initial detection operations thereof in that the interleaved loglikelihood ratios 814A and 814B are used in signal processing for the former. The inner MIMO detector 803 first calculates E(b0, b1, b2, b3, b4, b5, b6, b7) in the same manner as for initial detection. In addition, the coefficients corresponding to formula 11 and formula 32 are computed from the interleaved loglikelihood ratios 814A and 814B. The value of E(b0, b1, b2, b3, b4, b5, b6, b7) is corrected using the coefficients so calculated to obtain E′(b0, b1, b2, b3, b4, b5, b6, b7), which is output as the signal 804.
Loglikelihood calculator 805A takes the signal 804 as input, calculates the loglikelihood of bits b0, b1, b2, and b3, and outputs the loglikelihood signal 806A. Note that this loglikelihood calculation produces the loglikelihood of a bit being 1 and the loglikelihood of a bit being 0. The calculation scheme is as shown in formula 31 through formula 35, and the details are given by NonPatent Literature 2 and 3.
Similarly, loglikelihood calculator 805B takes the signal 804 as input, calculates the loglikelihood of bits b4, b5, b6, and b7, and outputs the loglikelihood signal 806A. Operations performed by the deinterleaver onwards are similar to those performed for initial detection.
While
The key point for the present embodiment is the calculation of H(t)×Y(t)×F. As shown in NonPatent Literature 5 and the like, QR decomposition may also be used to perform initial detection and iterative detection.
Also, as indicated by NonPatent Literature 11, MMSE (Minimum MeanSquare Error) and ZF (ZeroForcing) linear operations may be performed based on H(t)×Y(t)×F when performing initial detection.
As described above, when a transmission device according to the present embodiment using a MIMO system transmits a plurality of modulated signals from a plurality of antennas, changing the phase over time while multiplying by the precoding matrix so as to regularly change the phase results in improvements to data reception quality for a reception device in a LOS environment where direct waves are dominant, in contrast to a conventional spatial multiplexing MIMO system.
In the present embodiment, and particularly in the configuration of the reception device, the number of antennas is limited and explanations are given accordingly. However, the Embodiment may also be applied to a greater number of antennas. In other words, the number of antennas in the reception device does not affect the operations or advantageous effects of the present embodiment.
Also, although LDPC codes are described as a particular example, the present embodiment is not limited in this manner. Furthermore, the decoding scheme is not limited to the sumproduct decoding example given for the softin/softout decoder. Other softin/softout decoding schemes, such as the BCJR algorithm, SOVA, and the MaxLogMap algorithm may also be used. Details are provided in NonPatent Literature 6.
In addition, although the present embodiment is described using a singlecarrier scheme, no limitation is intended in this regard. The present embodiment is also applicable to multicarrier transmission. Accordingly, the present embodiment may also be realized using, for example, spreadspectrum communications, OFDM (Orthogonal FrequencyDivision Multiplexing), SCFDMA (Single Carrier FrequencyDivision Multiple Access), SCOFDM (Single Carrier Orthogonal FrequencyDivision Multiplexing), wavelet OFDM as described in NonPatent Literature 7, and so on. Furthermore, in the present embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner.
The following describes an example in which OFDM is used as a multicarrier scheme.
OFDMrelated processor 1201A takes weighted signal 309A as input, performs OFDMrelated processing thereon, and outputs transmit signal 1202A. Similarly, OFDMrelated processor 1201B takes postphasechange signal 309B as input, performs OFDMrelated processing thereon, and outputs transmit signal 1202A
Serialtoparallel converter 1302A performs serialtoparallel conversion on weighted signal 1301A (corresponding to weighted signal 309A from
Reorderer 1304A takes parallel signal 1303A as input, performs reordering thereof, and outputs reordered signal 1305A. Reordering is described in detail later.
IFFT (Inverse Fast Fourier Transform) unit 1306A takes reordered signal 1305A as input, applies an IFFT thereto, and outputs postIFFT signal 1307A.
Wireless unit 1308A takes postIFFT signal 1307A as input, performs processing such as frequency conversion and amplification, thereon, and outputs modulated signal 1309A. Modulated signal 1309A is then output as radio waves by antenna 1310A.
Serialtoparallel converter 1302B performs serialtoparallel conversion on weighted signal 1301B (corresponding to postphasechange signal 309B from
Reorderer 1304B takes parallel signal 1303B as input, performs reordering thereof, and outputs reordered signal 1305B. Reordering is described in detail later.
IFFT unit 1306B takes reordered signal 1305B as input, applies an IFFT thereto, and outputs postIFFT signal 1307B.
Wireless unit 1308B takes postIFFT signal 1307B as input, performs processing such as frequency conversion and amplification thereon, and outputs modulated signal 1309B. Modulated signal 1309B is then output as radio waves by antenna 1310A.
The transmission device from
As shown in
Similarly, with respect to the symbols of weighted signal 1301B input to serialtoparallel converter 1302B, the assigned ordering is #0, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, a different change of phase is applied to each of #0, #1, #2, and #3, which are equivalent to one period (cycle). Similarly, a different change of phase is applied to each of #4n, #4n+1, #4n+2, and #4n+3 (n being a nonzero positive integer), which are also equivalent to one period (cycle)
As shown in
The symbol group 1402 shown in
In the present embodiment, modulated signal z1 shown in
As such, when using a multicarrier transmission scheme such as OFDM, and unlike single carrier transmission, symbols may be arranged with respect to the frequency domain. Of course, the symbol arrangement scheme is not limited to those illustrated by
While
In
Here, symbol #0 is obtained through a change of phase at time u, symbol #1 is obtained through a change of phase at time u+1, symbol #2 is obtained through a change of phase at time u+2, and symbol #3 is obtained through a change of phase at time u+3.
Similarly, for frequencydomain symbol group 2220, symbol #4 is obtained through a change of phase at time u, symbol #5 is obtained through a change of phase at time u+1, symbol #6 is obtained through a change of phase at time u+2, and symbol #7 is obtained through a change of phase at time u+3.
The abovedescribed change of phase is applied to the symbol at time $1. However, in order to apply periodic shifting in the time domain, the following phase changes are applied to symbol groups 2201, 2202, 2203, and 2204.
For timedomain symbol group 2201, symbol #0 is obtained through a change of phase at time u, symbol #9 is obtained through a change of phase at time u+1, symbol #18 is obtained through a change of phase at time u+2, and symbol #27 is obtained through a change of phase at time u+3.
For timedomain symbol group 2202, symbol #28 is obtained through a change of phase at time u, symbol #1 is obtained through a change of phase at time u+1, symbol #10 is obtained through a change of phase at time u+2, and symbol #19 is obtained through a change of phase at time u+3.
For timedomain symbol group 2203, symbol #20 is obtained through a change of phase at time u, symbol #29 is obtained through a change of phase at time u+1, symbol #2 is obtained through a change of phase at time u+2, and symbol #11 is obtained through a change of phase at time u+3.
For timedomain symbol group 2204, symbol #12 is obtained through a change of phase at time u, symbol #21 is obtained through a change of phase at time u+1, symbol #30 is obtained through a change of phase at time u+2, and symbol #3 is obtained through a change of phase at time u+3.
The characteristic feature of
Although
In Embodiment 1, described above, phase changing is applied to a weighted (precoded with a fixed precoding matrix) signal z(t). The following Embodiments describe various phase changing schemes by which the effects of Embodiment 1 may be obtained.
In the abovedescribed Embodiment, as shown in
However, phase changing may also be applied before precoding is performed by the weighting unit 600. In addition to the components illustrated in
In such circumstances, the following configuration is possible. The phase changer 317B performs a regular change of phase with respect to baseband signal s2(t), on which mapping has been performed according to a selected modulation scheme, and outputs s2′(t)=s2(t)y(t) (where y(t) varies over time t). The weighting unit 600 executes precoding on s2′t, outputs z2(t)=W2s2′(t) (see formula 42) and the result is then transmitted.
Alternatively, phase changing may be performed on both modulated signals s_{1}(t) and s2(t). As such, the transmission device is configured so as to include a phase changer taking both signals output by the weighting unit 600, as shown in
Like phase changer 317B, phase changer 317A performs regular a regular change of phase on the signal input thereto, and as such changes the phase of signal z1′(t) precoded by the weighting unit. Postphasechange signal z1(t) is then output to a transmitter.
However, the phase changing rate applied by the phase changers 317A and 317B varies simultaneously in order to perform the phase changing shown in
Also, as described above, a change of phase may be performed before precoding is performed by the weighting unit. In such a case, the transmission device should be configured as illustrated in
When a change of phase is carried out on both modulated signals, each of the transmit signals is, for example, control information that includes information about the phase changing pattern. By obtaining the control information, the reception device knows the phase changing scheme by which the transmission device regularly varies the change, i.e., the phase changing pattern, and is thus able to demodulate (decode) the signals correctly.
Next, variants of the sample configurations shown in
Phase changer 317A of
Here, a change of phase having a period (cycle) of four is, for example, applied to z1′(t). (Meanwhile, the phase of z2′(t) is not changed.) Accordingly, for time u, y_{1}(u)=e^{j0 }and y_{2}(u)=1, for time u+1, y_{1}(u+1)=e^{jπ/2 }and y_{2}(u+1)=1, for time u+2, y_{1}(u+2)=e^{jπ} and y_{2}(u+2)=1, and for time u+3, y_{1}(u+3)=e^{j3π/2 }and y_{2}(u+3)=1.
Next, a change of phase having a period (cycle) of four is, for example, applied to z2′(t). (Meanwhile, the phase of z1′(t) is not changed.) Accordingly, for time u+4, y_{1}(u+4)=1 and y_{2}(u+4)=e^{j0}, for time u+5, y_{1}(u+5)=1 and y_{2}(u+5)=e^{jπ/2}, for time u+6, y_{1}(u+6)=1 and y_{2}(u+6)=e^{jπ}, and for time u+7, y_{1}(u+7)=1 and y_{2}(u+7)=e^{j3π/2}.
Accordingly, given the above examples.
for any time 8k, y_{1}(8k)=e^{j0 }and y_{2}(8k)=1,
for any time 8k+1, y_{1}(8k+1)=e^{jπ/2 }and y_{2}(8k+1)=1,
for any time 8k+2, y_{1}(8k+2)=e^{jπ} and y_{2}(8k+2)=1,
for any time 8k+3, y_{1}(8k+3)=e^{j3π/2 }and y_{2}(8k+3)=1,
for any time 8k+4, y_{1}(8k+4)=1 and y_{2}(8k+4)=e^{j0},
for any time 8k+5, y_{1}(8k+3)=1 and y_{2}(8k+5)=e^{jπ/2 }
for any time 8k+6, y_{1}(8k+6)=1 and y_{2}(8k+6)=e^{jπ}, and
for any time 8k+7, y_{1}(8k+7)=1 and y_{2}(8k+7)=e^{j3π/2}.
As described above, there are two intervals, one where the change of phase is performed on z1′(t) only, and one where the change of phase is performed on z2′(t) only. Furthermore, the two intervals form a phase changing period (cycle). While the above explanation describes the interval where the change of phase is performed on z1′(t) only and the interval where the change of phase is performed on z2′(t) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing a change of phase having a period (cycle) of four on z1′(t) only and then performing a change of phase having a period (cycle) of four on z2′(t) only, no limitation is intended in this manner. The changes of phase may be performed on z1′(t) and on z2′(t) in any order (e.g., the change of phase may alternate between being performed on z1′(t) and on z2′(t), or may be performed in random order).
Phase changer 317A of
Here, a change of phase having a period (cycle) of four is, for example, applied to s1(t). (Meanwhile, s2(t) remains unchanged). Accordingly, for time u, y_{1}(u)=e^{j0 }and y_{2}(u)=1, for time u+1, y_{1}(u+1)=e^{jπ/2 }and y_{2}(u+1)=1, for time u+2, y_{1}(u+2)=e^{j0 }and y_{2}(u+2)=1, and for time u+3, y_{1}(u+3)=e^{j3π/2 }and y_{2}(u+3)=1.
Next, a change of phase having a period (cycle) of four is, for example, applied to s2(t). (Meanwhile, s1(t) remains unchanged). Accordingly, for time u+4, y_{1}(u+4)=1 and y_{2}(u+4)=e^{j0}, for time u+5, y_{1}(u+5)=1 and y_{2}(u+5)=e^{jπ/2}, for time u+6, y_{1}(u+6)=1 and y_{2}(u+6)=e^{jπ}, and for time u+7, y_{1}(u+7)=1 and y_{2}(u+7)=e^{j3π/2}.
Accordingly, given the above examples,
for any time 8k, y_{1}(8k)=e^{j0 }and y_{2}(8k)=1,
for any time 8k+1, y_{1}(8k+1)=e^{jπ/2 }and y_{2}(8k+1)=1,
for any time 8k+2, y_{1}(8k+2)=e^{jπ} and y_{2}(8k+2)=1,
for any time 8k+3, y_{1}(8k+3)=e^{j3π/2 }and y_{2}(8k+3)=1,
for any time 8k+4, y_{1}(8k+4)=1 and y_{2}(8k+4)=e^{j0},
for any time 8k+5, y_{1}(8k+5)=1 and y_{2}(8k+5)=e^{jπ/2 }
for any time 8k+6, y_{1}(8k+6)=1 and y_{2}(8k+6)=e^{jπ}, and
for any time 8k+7, y_{1}(8k+7)=1 and y_{2}(8k+7)=e^{j3π/2}.
As described above, there are two intervals, one where the change of phase is performed on s_{1}(t) only, and one where the change of phase is performed on s2(t) only. Furthermore, the two intervals form a phase changing period (cycle). Although the above explanation describes the interval where the change of phase is performed on s1(t) only and the interval where the change of phase is performed on s2(t) only as being equal, no limitation is intended in this manner. The two intervals may also differ. In addition, while the above explanation describes performing the change of phase having a period (cycle) of four on s1(t) only and then performing the change of phase having a period (cycle) of four on s2(t) only, no limitation is intended in this manner. The changes of phase may be performed on s1(t) and on s2(t) in any order (e.g., may alternate between being performed on s1(t) and on s2(t), or may be performed in random order).
Accordingly, the reception conditions under which the reception device receives each transmit signal z1(t) and z2(t) are equalized. By periodically switching the phase of the symbols in the received signals z1(t) and z2(t), the ability of the error corrected codes to correct errors may be improved, thus ameliorating received signal quality in the LOS environment.
Accordingly, Embodiment 2 as described above is able to produce the same results as the previously described Embodiment 1.
Although the present embodiment used a singlecarrier scheme, i.e., time domain phase changing, as an example, no limitation is intended in this regard. The same effects are also achievable using multicarrier transmission. Accordingly, the present embodiment may also be realized using, for example, spreadspectrum communications, OFDM, SCFDMA (Single Carrier FrequencyDivision Multiple Access), SCOFDM, wavelet OFDM as described in NonPatent Literature 7, and so on. As previously described, while the present embodiment explains the change of phase as changing the phase with respect to the time domain t, the phase may alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the phase changing scheme in the time domain t described in the present embodiment and replacing t with f (f being the ((sub) carrier) frequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing scheme of the present embodiment is also applicable to changing the phase with respect both the time domain and the frequency domain.
Accordingly, although
Furthermore, in the present embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner.
Embodiments 1 and 2, described above, discuss regular changes of phase. Embodiment 3 describes a scheme of allowing the reception device to obtain good received signal quality for data, regardless of the reception device arrangement, by considering the location of the reception device with respect to the transmission device.
Embodiment 3 concerns the symbol arrangement within signals obtained through a change of phase.
First, an example is explained in which the change of phase is performed one of two baseband signals, precoded as explained in Embodiment 1 (see
(Although
Consider symbol 3100 at carrier 2 and time $2 of
Within carrier 2, there is a very strong correlation between the channel conditions for symbol 3100 at carrier 2, time $2 and the channel conditions for the time domain nearestneighbour symbols to time $2, i.e., symbol 3013 at time $1 and symbol 3101 at time $3 within carrier 2.
Similarly, for time $2, there is a very strong correlation between the channel conditions for symbol 3100 at carrier 2, time $2 and the channel conditions for the frequencydomain nearestneighbour symbols to carrier 2, i.e., symbol 3104 at carrier 1, time $2 and symbol 3104 at time $2, carrier 3.
As described above, there is a very strong correlation between the channel conditions for symbol 3100 and the channel conditions for symbols 3101, 3102, 3103, and 3104.
The present description considers N different phases (N being an integer, N≥2) for multiplication in a transmission scheme where the phase is regularly changed. The symbols illustrated in
The present embodiment takes advantage of the high correlation in channel conditions existing between neighbouring symbols in the frequency domain and/or neighbouring symbols in the time domain in a symbol arrangement enabling high data reception quality to be obtained by the reception device receiving the phasechanged symbols.
In order to achieve this high data reception quality, conditions #1 and #2 are necessary.
As shown in
As shown in
Ideally, data symbols satisfying Condition #1 should be present. Similarly, data symbols satisfying Condition #2 should be present.
The reasons supporting Conditions #1 and #2 are as follows.
A very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the time domain, as described above.
Accordingly, when three neighbouring symbols in the time domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding.
Similarly, a very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the frequency domain, as described above.
Accordingly, when three neighbouring symbols in the frequency domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding.
Combining Conditions #1 and #2, ever greater data reception quality is likely achievable for the reception device. Accordingly, the following Condition #3 can be derived.
As shown in
Here, the different changes in phase are as follows. Changes in phase are defined from 0 radians to 271 radians. For example, for time X, carrier Y, a phase change of e^{jθX,Y }is applied to precoded baseband signal z2′ from
Ideally, a data symbol should satisfy Condition #3.
As evident from
In other words, in
Similarly, in
Similarly, in
The following describes an example in which a change of phase is performed on two precoded baseband signals, as explained in Embodiment 2 (see
When a change of phase is performed on precoded baseband signal z1′ and precoded baseband signal z2′ as shown in
Scheme 1 involves a change in phase performed on precoded baseband signal z2′ as described above, to achieve the change in phase illustrated by
The symbols illustrated in
As shown in
As described above, the change in phase performed on precoded baseband signal z2′ has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the change in phase applied to precoded baseband signal z1′ and to precoded baseband signal z2′ into consideration. Accordingly, data reception quality may be improved for the reception device.
Scheme 2 involves a change in phase of precoded baseband signal z2′ as described above, to achieve the change in phase illustrated by
The symbols illustrated in
As described above, the change in phase performed on precoded baseband signal z2′ has a period (cycle) of ten, but by taking the changes in phase applied to precoded baseband signal z1′ and precoded baseband signal z2′ into consideration, the period (cycle) can be effectively made equivalent to 30 for both precoded baseband signals z1′ and z2′. Accordingly, data reception quality may be improved for the reception device. An effective way of applying scheme 2 is to perform a change in phase on precoded baseband signal z1′ with a period (cycle) of N and perform a change in phase on precoded baseband signal z2′ with a period (cycle) of M such that N and M are coprime. As such, by taking both precoded baseband signals z1′ and z2′ into consideration, a period (cycle) of NxM is easily achievable, effectively making the period (cycle) greater when N and M are coprime.
The above describes an example of the phase changing scheme pertaining to Embodiment 3. The present invention is not limited in this manner. As explained for Embodiments 1 and 2, a change in phase may be performed with respect the frequency domain or the time domain, or on timefrequency blocks. Similar improvement to the data reception quality can be obtained for the reception device in all cases.
The same also applies to frames having a configuration other than that described above, where pilot symbols (SP (Scattered Pilot)) and symbols transmitting control information are inserted among the data symbols. The details of change in phase in such circumstances are as follows.
The key point of
The key point of
The key point of
The key point of
In
In
Although not indicated in the frame configurations from
Wireless units 310A and 310B of
A selector 5301 takes the plurality of baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal 313 for output.
Similarly, as shown in
The above explanations are given using pilot symbols, control symbols, and data symbols as examples. However, the present invention is not limited in this manner. When symbols are transmitted using schemes other than precoding, such as singleantenna transmission or transmission using spacetime block coding, not performing a change of phase is important. Conversely, performing a change of phase on symbols that have been precoded is the key point of the present invention.
Accordingly, a characteristic feature of the present invention is that the change of phase is not performed on all symbols within the frame configuration in the timefrequency domain, but only performed on signals that have been precoded.
Embodiments 1 and 2, described above, discuss a regular change of phase. Embodiment 3, however, discloses performing a different change of phase on neighbouring symbols.
The present embodiment describes a phase changing scheme that varies according to the modulation scheme and the coding rate of the errorcorrecting codes used by the transmission device.
Table 1, below, is a list of phase changing scheme settings corresponding to the settings and parameters of the transmission device.
In Table 1, #1 denotes modulated signal s1 from Embodiment 1 described above (baseband signal s1 modulated with the modulation scheme set by the transmission device) and #2 denotes modulated signal s2 (baseband signal s2 modulated with the modulation scheme set by the transmission device). The coding rate column of Table 1 indicates the coding rate of the errorcorrecting codes for modulation schemes #1 and #2. The phase changing pattern column of Table 1 indicates the phase changing scheme applied to precoded baseband signals z1 (z1′) and z2 (z2′), as explained in Embodiments 1 through 3. Although the phase changing patterns are labeled A, B, C, D, E, and so on, this refers to the phase change degree applied, for example, in a phase changing pattern given by formula 46 and formula 47, above. In the phase changing pattern column of Table 1, the dash signifies that no change of phase is applied.
The combinations of modulation scheme and coding rate listed in Table 1 are examples. Other modulation schemes (such as 128QAM and 256QAM) and coding rates (such as 7/8) not listed in Table 1 may also be included. Also, as described in Embodiment 1, the errorcorrecting codes used for s1 and s2 may differ (Table 1 is given for cases where a single type of errorcorrecting codes is used, as in
In Embodiments 1 through 3, the change of phase is applied to precoded baseband signals. However, the amplitude may also be modified along with the phase in order to apply periodical, regular changes. Accordingly, an amplification modification pattern regularly modifying the amplitude of the modulated signals may also be made to conform to Table 1. In such circumstances, the transmission device should include an amplification modifier that modifies the amplification after weighting unit 308A or weighting unit 308B from
Furthermore, although not indicated in Table 1 above, the mapping scheme may also be regularly modified by the mapper, without a regular change of phase.
That is, when the mapping scheme for modulated signal s1(t) is 16QAM and the mapping scheme for modulated signal s2(t) is also 16QAM, the mapping scheme applied to modulated signal s2(t) may be regularly changed as follows: from 16QAM to 16APSK, to 16QAM in the I (inphase)Q (quadrature(phase)) plane, to a first mapping scheme producing a signal point arrangement (constellation) unlike 16APSK, to 16QAM in the I (inphase)Q (quadrature(phase)) plane, to a second mapping scheme producing a signal point arrangement (constellation) unlike 16APSK, and so on. As such, the data reception quality can be improved for the reception device, much like the results obtained by a regular change of phase described above.
In addition, the present invention may use any combination of schemes for a regular change of phase, mapping scheme, and amplitude, and the transmit signal may transmit with all of these taken into consideration.
The present embodiment may be realized using singlecarrier schemes as well as multicarrier schemes. Accordingly, the present embodiment may also be realized using, for example, spreadspectrum communications, OFDM, SCFDMA, SCOFDM, wavelet OFDM as described in NonPatent Literature 7, and so on. As described above, the present embodiment describes changing the phase, amplitude, and mapping schemes by performing phase, amplitude, and mapping scheme modifications with respect to the time domain t. However, much like Embodiment 1, the same changes may be carried out with respect to the frequency domain. That is, considering the phase, amplitude, and mapping scheme modification in the time domain t described in the present embodiment and replacing t with f (f being the ((sub) carrier) frequency) leads to phase, amplitude, and mapping scheme modification applicable to the frequency domain. Also, the phase, amplitude, and mapping scheme modification of the present embodiment is also applicable to phase, amplitude, and mapping scheme modification in both the time domain and the frequency domain.
Furthermore, in the present embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, etc) or symbols transmitting control information, may be arranged within the frame in any manner.
The present embodiment describes a scheme for regularly changing the phase when encoding is performed using block codes as described in NonPatent Literature 12 through 15, such as QC (QuasiCyclic) LDPC Codes (not only QCLDPC but also LDPC codes may be used), concatenated LDPC and BCH (BoseChaudhuriHocquenghem) codes, Turbo codes or DuoBinary Turbo Codes using tailbiting, and so on. The following example considers a case where two streams s1 and s2 are transmitted. However, when encoding has been performed using block codes and control information and the like is not required, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC (cyclic redundancy check) transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information.
As shown in
Then, given that the transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 750 slots are needed to transmit all of the bits making up a single coded block, and when the modulation scheme is 64QAM, 500 slots are needed to transmit all of the bits making up a single coded block.
The following describes the relationship between the abovedefined slots and the phase of multiplication, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase. That is, five different phase changing values (or phase changing sets) have been prepared for the phase changer of the transmission device from
For the abovedescribed 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality.
Similarly, for the abovedescribed 700 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 150 slots, PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, and PHASE[4] is used on 150 slots.
Furthermore, for the abovedescribed 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64QAM, PHASE[0] is used on 100 slots, PHASE[1] is used on 100 slots, PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, and PHASE[4] is used on 100 slots.
As described above, a scheme for a regular change of phase requires the preparation of N phase changing values (or phase changing sets) (where the N different phases are expressed as PHASE[0], PHASE[1], PHASE[2], PHASE[N2], PHASE[N−1]). As such, in order to transmit all of the bits making up a single coded block, PHASE[0] is used on K_{0 }slots, PHASE[1] is used on K_{1 }slots, PHASE[i] is used on K_{i }slots (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1] is used on K_{N−1 }slots, such that Condition # A01 is met.
K_{0}=K_{1 }. . . =K_{i}=K_{N−1}. That is, K_{a}=K_{b }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a b).
Then, when a communication system that supports multiple modulation schemes selects one such supported modulation scheme for use, Condition # A01 is preferably satisfied for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition # A01 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition # A01.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}−K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
As shown in
The transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 1500 slots are needed to transmit all of the bits making up the two coded blocks, and when the modulation scheme is 64QAM, 1000 slots are needed to transmit all of the bits making up the two coded blocks.
The following describes the relationship between the abovedefined slots and the phase of multiplication, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase. That is, five different phase changing values (or phase changing sets) have been prepared for the phase changers of the transmission devices from
For the abovedescribed 3000 slots needed to transmit the 6000×2 bits making up a single coded block when the modulation scheme is QPSK, PHASE[0] is used on 600 slots, PHASE[1] is used on 600 slots, PHASE[2] is used on 600 slots, PHASE[3] is used on 600 slots, and PHASE[4] is used on 600 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality.
Furthermore, in order to transmit the first coded block, PHASE[0] is used on slots 600 times, PHASE[1] is used on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] is used on slots 600 times. Furthermore, in order to transmit the second coded block, PHASE[0] is used on slots 600 times, PHASE[1] is used on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] is used on slots 600 times.
Similarly, for the abovedescribed 1500 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 16QAM, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots.
Furthermore, in order to transmit the first coded block, PHASE[0] is used on slots 300 times, PHASE[1] is used on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] is used on slots 300 times. Furthermore, in order to transmit the second coded block, PHASE[0] is used on slots 300 times, PHASE[1] is used on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] is used on slots 300 times.
Similarly, for the abovedescribed 1000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 64QAM, PHASE[0] is used on 200 slots, PHASE[1] is used on 200 slots, PHASE[2] is used on 200 slots, PHASE[3] is used on 200 slots, and PHASE[4] is used on 200 slots.
Furthermore, in order to transmit the first coded block, PHASE[0] is used on slots 200 times, PHASE[1] is used on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] is used on slots 200 times. Furthermore, in order to transmit the second coded block, PHASE[0] is used on slots 200 times, PHASE[1] is used on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] is used on slots 200 times.
As described above, a scheme for regularly changing the phase requires the preparation of phase changing values (or phase changing sets) expressed as PHASE[0], PHASE[1], PHASE[2], . . . , PHASE[N−2], PHASE[N−1]. As such, in order to transmit all of the bits making up two coded blocks, PHASE[0] is used on K_{0 }slots, PHASE[1] is used on K_{1 }slots, PHASE[i] is used on K_{i }slots (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1), and PHASE[N−1] is used on K_{N−1 }slots, such that Condition # A03 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K_{N−1}. That is, K_{a}=K_{b }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1),a b).
Further, in order to transmit all of the bits making up the first coded block, PHASE[0] is used K_{0,1 }times, PHASE[1] is used K_{1,1 }times, PHASE[i] is used K_{i,1 }times (where i=0, 1, 2, . . . , N−1(i denotes an integer that satisfies 0≤i≤N−1), and PHASE[N−1] is used K_{N−1,1 }times, such that Condition # A04 is met.
K_{0,1}=K_{1,1}= . . . K_{i,i}= . . . K_{N−1,1}. That is, K_{a,1}=K_{b,1 }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Furthermore, in order to transmit all of the bits making up the second coded block, PHASE[0] is used K_{0,2 }times, PHASE[1] is used K_{1,2 }times, PHASE[i] is used K_{i,2 }times (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1), and PHASE[N−1] is used K_{N−1,2 }times, such that Condition # A05 is met.
K_{0,2}=K_{1,2}= . . . K_{i,2}= . . . K_{N−1,2}. That is, K_{a,2}=K_{b,2 }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Then, when a communication system that supports multiple modulation schemes selects one such supported modulation scheme for use, Condition # A03, # A04, and # A05 should preferably be met for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbol (though some may happen to use the same number), Conditions # A03, # A04, and # A05 may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition # A03, # A04, and # A05.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}−K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
The difference between K_{a,1 }and K_{b,1 }satisfies 0 or 1. That is, K_{a,1}−K_{b,1} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1, (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1) a≠b)
The difference between K_{a,2 }and K_{b,2 }satisfies 0 or 1. That is, K_{a,2}−K_{b,2} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
As described above, bias among the phases being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase of multiplication. As such, data reception quality can be improved for the reception device.
In the present embodiment N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the scheme for a regular change of phase. As such, N phase changing values (or phase changing sets) PHASE[0], PHASE[1], PHASE[2], . . . , PHASE[N−2], and PHASE[N−1] are prepared. However, schemes exist for reordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) may also change the phases of blocks in the time domain or in the timefrequency domain to obtain a symbol arrangement as described in Embodiment 1. Although the above examples discuss a phase changing scheme with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always for a regular period (cycle). As long as the abovedescribed conditions are satisfied, great quality data reception improvements are realizable for the reception device.
Furthermore, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase (the transmission schemes described in Embodiments 1 through 4), the transmission device (broadcaster, base station) may select any one of these transmission schemes.
As described in NonPatent Literature 3, spatial multiplexing MIMO schemes involve transmitting signals s1 and s2, which are mapped using a selected modulation scheme, on each of two different antennas. As described in Embodiments 1 through 4, MIMO schemes using a fixed precoding matrix involve performing precoding only (with no change of phase). Further, spacetime block coding schemes are described in NonPatent Literature 9, 16, and 17. Singlestream transmission schemes involve transmitting signal s1, mapped with a selected modulation scheme, from an antenna after performing predetermined processing.
Schemes using multicarrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multicarrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub)carrier group are preferably used to realize the present embodiment.
When a change of phase is performed, then for example, a phase changing value for PHASE[i] of X radians is performed on only one precoded baseband signal, the phase changers of
The following describes a sample configuration of an application of the transmission schemes and reception schemes discussed in the above embodiments and a system using the application.
The signals transmitted by the broadcaster 3601 are received by an antenna (such as antenna 3660 or 3640) embedded within or externally connected to each of the receivers. Each receiver obtains the multiplexed data by using reception schemes discussed in the abovedescribed Embodiments to demodulate the signals received by the antenna. Accordingly, the digital broadcasting system 3600 is able to realize the effects of the present invention, as discussed in the abovedescribed Embodiments.
The video data included in the multiplexed data are coded with a video coding method compliant with a standard such as MPEG2 (Moving Picture Experts Group), MPEG4AVC (Advanced Video Coding), VC1, or the like. The audio data included in the multiplexed data are encoded with an audio coding method compliant with a standard such as Dolby AC3 (Audio Coding), Dolby Digital Plus, MLP (Meridian Lossless Packing), DTS (Digital Theater Systems), DTSHD, PCM (PulseCode Modulation), or the like.
The receiver 3700 further includes a stream interface 3720 that demultiplexes the audio and video data in the multiplexed data obtained by the demodulator 3702, a signal processor 3704 that decodes the video data obtained from the demultiplexed video data into a video signal by applying a video decoding method corresponding thereto and decodes the audio data obtained from the demultiplexed audio data into an audio signal by applying an audio decoding method corresponding thereto, an audio output unit 3706 that outputs the decoded audio signal through a speaker or the like, and a video display unit 3707 that outputs the decoded video signal on a display or the like.
When, for example, a user uses a remote control 3750, information for a selected channel (selected (television) program or audio broadcast) is transmitted to an operation input unit 3710. Then, the receiver 3700 performs processing on the received signal received by the antenna 3760 that includes demodulating the signal corresponding to the selected channel, performing errorcorrecting decoding, and so on, in order to obtain the received data. At this point, the receiver 3700 obtains control symbol information that includes information on the transmission scheme (the transmission scheme, modulation scheme, errorcorrection scheme, and so on from the abovedescribed Embodiments) (as described using
According to this configuration, the user is able to view programs received by the receiver 3700.
The receiver 3700 pertaining to the present embodiment further includes a drive 3708 that may be a magnetic disk, an optical disc, a nonvolatile semiconductor memory, or a similar recording medium. The receiver 3700 stores data included in the demultiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding (in some circumstances, the data obtained through demodulation by the demodulator 3702 may not be subject to error correction. Also, the receiver 3700 may perform further processing after error correction. The same hereinafter applies to similar statements concerning other components), data corresponding to such data (e.g., data obtained through compression of such data), data obtained through audio and video processing, and so on, on the drive 3708. Here, an optical disc is a recording medium, such as DVD (Digital Versatile Disc) or BD™ (Bluray Disc), that is readable and writable with the use of a laser beam. A magnetic disk is a floppy disk, a hard disk, or similar recording medium on which information is storable through the use of magnetic flux to magnetize a magnetic body. A nonvolatile semiconductor memory is a recording medium, such as flash memory or ferroelectric random access memory, composed of semiconductor element(s). Specific examples of nonvolatile semiconductor memory include an SD card using flash memory and a Flash SSD (Solid State Drive). Naturally, the specific types of recording media mentioned herein are merely examples. Other types of recording mediums may also be used.
According to this structure, the user is able to record and store programs received by the receiver 3700, and is thereby able to view programs at any given time after broadcasting by reading out the recorded data thereof.
Although the above explanations describe the receiver 3700 storing multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding on the drive 3708, a portion of the data included in the multiplexed data may instead be extracted and recorded. For example, when data broadcasting services or similar content is included along with the audio and video data in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, the audio and video data may be extracted from the multiplexed data demodulated by the demodulator 3702 and stored as new multiplexed data. Furthermore, the drive 3708 may store either the audio data or the video data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding as new multiplexed data. The aforementioned data broadcasting service content included in the multiplexed data may also be stored on the drive 3708.
Furthermore, when a television, recording device (e.g., a DVD recorder, BD recorder HDD recorder, SD card, or similar), or mobile phone incorporating the receiver 3700 of the present invention receives multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding that includes data for correcting bugs in software used to operate the television or recording device, for correcting bugs in software for preventing personal information and recorded data from being leaked, and so on, such software bugs may be corrected by installing the data on the television or recording device. As such, bugs in the receiver 3700 are corrected through the inclusion of data for correcting bugs in the software of the receiver 3700. Accordingly, the television, recording device, or mobile phone incorporating the receiver 3700 may be made to operate more reliably.
Here, the process of extracting a portion of the data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding is performed by, for example, the stream interface 3703. Specifically, the stream interface 3703, demultiplexes the various data included in the multiplexed data demodulated by the demodulator 3702, such as audio data, video data, data broadcasting service content, and so on, as instructed by a nondiagrammed controller such as a CPU. The stream interface 3703 then extracts and multiplexes only the indicated demultiplexed data, thus generating new multiplexed data. The data to be extracted from the demultiplexed data may be determined by the user or may be determined in advance according to the type of recording medium.
According to such a structure, the receiver 3700 is able to extract and record only the data needed in order to view the recorded program. As such, the amount of data to be recorded can be reduced.
Although the above explanation describes the drive 3708 as storing multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, the video data included in the multiplexed data so obtained may be converted by using a different video coding method than the original video coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The drive 3708 may then store the converted video data as new multiplexed data. Here, the video coding method used to generate the new video data may conform to a different standard than that used to generate the original video data. Alternatively, the same video coding method may be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding may be converted by using a different audio coding method than the original audio coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The drive 3708 may then store the converted audio data as new multiplexed data.
Here, the process by which the audio or video data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding is converted so as to reduce the amount of data or the bit rate thereof is performed by, for example, the stream interface 3703 or the signal processor 3704. Specifically, the stream interface 3703 demultiplexes the various data included in the multiplexed data demodulated by the demodulator 3702, such as audio data, video data, data broadcasting service content, and so on, as instructed by an undiagrammed controller such as a CPU. The signal processor 3704 then performs processing to convert the video data so demultiplexed by using a different video coding method than the original video coding method applied thereto, and performs processing to convert the audio data so demultiplexed by using a different video coding method than the original audio coding method applied thereto. As instructed by the controller, the stream interface 3703 then multiplexes the converted audio and video data, thus generating new multiplexed data. The signal processor 3704 may, in accordance with instructions from the controller, performing conversion processing on either the video data or the audio data, alone, or may perform conversion processing on both types of data. In addition, the amounts of video data and audio data or the bit rate thereof to be obtained by conversion may be specified by the user or determined in advance according to the type of recording medium.
According to such a structure, the receiver 3700 is able to modify the amount of data or the bitrate of the audio and video data for storage according to the data storage capacity of the recording medium, or according to the data reading or writing speed of the drive 3708. Therefore, programs can be stored on the drive despite the storage capacity of the recording medium being less than the amount of multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, or the data reading or writing speed of the drive being lower than the bit rate of the demultiplexed data obtained through demodulation by the demodulator 3702. As such, the user is able to view programs at any given time after broadcasting by reading out the recorded data.
The receiver 3700 further includes a stream output interface 3709 that transmits the multiplexed data demultiplexed by the demodulator 3702 to external devices through a communications medium 3730. The stream output interface 3709 may be, for example, a wireless communication device transmitting modulated multiplexed data to an external device using a wireless transmission scheme conforming to a wireless communication standard such as WiFi™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGig, WirelessHD, Bluetooth, ZigBee, and so on through a wireless medium (corresponding to the communications medium 3730). The stream output interface 3709 may also be a wired communication device transmitting modulated multiplexed data to an external device using a communication scheme conforming to a wired communication standard such as Ethernet™, USB (Universal Serial Bus), PLC (Power Line Communication), HDMI™ (HighDefinition Multimedia Interface) and so on through a wired transmission path (corresponding to the communications medium 3730) connected to the stream output interface 3709.
According to this configuration, the user is able to use an external device with the multiplexed data received by the receiver 3700 using the reception scheme described in the abovedescribed Embodiments. The usage of multiplexed data by the user here includes use of the multiplexed data for realtime viewing on an external device, recording of the multiplexed data by a recording unit included in an external device, and transmission of the multiplexed data from an external device to a yet another external device.
Although the above explanations describe the receiver 3700 outputting multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding through the stream output interface 3709, a portion of the data included in the multiplexed data may instead be extracted and output. For example, when data broadcasting services or similar content is included along with the audio and video data in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, the audio and video data may be extracted from the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, multiplexed and output by the stream output interface 3709 as new multiplexed data. In addition, the stream output interface 3709 may store either the audio data or the video data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding as new multiplexed data.
Here, the process of extracting a portion of the data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding is performed by, for example, the stream interface 3703. Specifically, the stream interface 3703 demultiplexes the various data included in the multiplexed data demodulated by the demodulator 3702, such as audio data, video data, data broadcasting service content, and so on, as instructed by an undiagrammed controller such as a CPU. The stream interface 3703 then extracts and multiplexes only the indicated demultiplexed data, thus generating new multiplexed data. The data to be extracted from the demultiplexed data may be determined by the user or may be determined in advance according to the type of stream output interface 3709.
According to this structure, the receiver 3700 is able to extract and output only the required data to an external device. As such, fewer multiplexed data are output using less communication band.
Although the above explanation describes the stream output interface 3709 as outputting multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, the video data included in the multiplexed data so obtained may be converted by using a different video coding method than the original video coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The stream output interface 3709 may then output the converted video data as new multiplexed data. Here, the video coding method used to generate the new video data may conform to a different standard than that used to generate the original video data. Alternatively, the same video coding method may be used with different parameters. Similarly, the audio data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding may be converted by using a different audio coding method than the original audio coding method applied thereto, so as to reduce the amount of data or the bit rate thereof. The stream output interface 3709 may then output the converted audio data as new multiplexed data.
Here, the process by which the audio or video data included in the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding is converted so as to reduce the amount of data or the bit rate thereof is performed by, for example, the stream interface 3703 or the signal processor 3704. Specifically, the stream interface 3703 demultiplexes the various data included in the multiplexed data demodulated by the demodulator 3702, such as audio data, video data, data broadcasting service content, and so on, as instructed by an undiagrammed controller. The signal processor 3704 then performs processing to convert the video data so demultiplexed by using a different video coding method than the original video coding method applied thereto, and performs processing to convert the audio data so demultiplexed by using a different video coding method than the original audio coding method applied thereto. As instructed by the controller, the stream interface 3703 then multiplexes the converted audio and video data, thus generating new multiplexed data. The signal processor 3704 may, in accordance with instructions from the controller, performing conversion processing on either the video data or the audio data, alone, or may perform conversion processing on both types of data. In addition, the amounts of video data and audio data or the bit rate thereof to be obtained by conversion may be specified by the user or determined in advance according to the type of stream output interface 3709.
According to this structure, the receiver 3700 is able to modify the bit rate of the video and audio data for output according to the speed of communication with the external device. Thus, despite the speed of communication with an external device being slower than the bit rate of the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding, by outputting new multiplexed data from the stream output interface to the external device, the user is able to use the new multiplexed data with other communication devices.
The receiver 3700 further includes an audiovisual output interface 3711 that outputs audio and video signals decoded by the signal processor 3704 to the external device through an external communications medium. The audiovisual output interface 3711 may be, for example, a wireless communication device transmitting modulated audiovisual data to an external device using a wireless transmission scheme conforming to a wireless communication standard such as WiFi™ (IEEE 802.11a, IEEE 802.11b, IEEE 802.11g, IEEE 802.11n, and so on), WiGig, WirelessHD, Bluetooth, ZigBee, and so on through a wireless medium. The stream output interface 3709 may also be a wired communication device transmitting modulated audiovisual data to an external device using a communication scheme conforming to a wired communication standard such as Ethernet™, USB, PLC, HDMI, and so on through a wired transmission path connected to the stream output interface 3709. Furthermore, the stream output interface 3709 may be a terminal for connecting a cable that outputs analogue audio signals and video signals asis.
According to such a structure, the user is able to use the audio signals and video signals decoded by the signal processor 3704 with an external device.
Further, the receiver 3700 includes an operation input unit 3710 that receives user operations as input. The receiver 3700 behaves in accordance with control signals input by the operation input unit 3710 according to user operations, such as by switching the power supply ON or OFF, changing the channel being received, switching subtitle display ON or OFF, switching between languages, changing the volume output by the audio output unit 3706, and various other operations, including modifying the settings for receivable channels and the like.
The receiver 3700 may further include functionality for displaying an antenna level representing the received signal quality while the receiver 3700 is receiving a signal. The antenna level may be, for example, a index displaying the received signal quality calculated according to the RSSI (Received Signal Strength Indicator), the received signal magnetic field strength, the C/N (carriertonoise) ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on, received by the receiver 3700 and indicating the level and the quality of a received signal. In such circumstances, the demodulator 3702 includes a signal quality calibrator that measures the RSSI, the received signal magnetic field strength, the C/N ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on. In response to user operations, the receiver 3700 displays the antenna level (signal level, signal quality) in a userrecognizable format on the video display unit 3707. The display format for the antenna level (signal level, signal quality) may be a numerical value displayed according to the RSSI, the received signal magnetic field strength, the C/N ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on, or may be an image display that varies according to the RSSI, the received signal magnetic field strength, the C/N ratio, the BER, the packet error rate, the frame error rate, the channel state information, and so on. The receiver 3700 may display multiple antenna level (signal level, signal quality) calculated for each stream s_{1}, s2, and so on demultiplexed using the reception scheme discussed in the abovedescribed Embodiments, or may display a single antenna level (signal level, signal quality) calculated for all such streams. When the video data and audio data composing a program are transmitted hierarchically, the signal level (signal quality) may also be displayed for each hierarchical level.
According to the above structure, the user is given an understanding of the antenna level (signal level, signal quality) numerically or visually during reception using the reception schemes discussed in the abovedescribed Embodiments.
Although the above example describes the receiver 3700 as including the audio output unit 3706, the video display unit 3707, the drive 3708, the stream output interface 3709, and the audiovisual output interface 3711, all of these components are not strictly necessary. As long as the receiver 3700 includes at least one of the abovedescribed components, the user is able to use the multiplexed data obtained through demodulation by the demodulator 3702 and errorcorrecting decoding. Any receiver may be freely combined with the abovedescribed components according to the usage scheme.
The following is a detailed description of a sample configuration of multiplexed data. The data configuration typically used in broadcasting is an MPEG2 transport stream (TS). Therefore the following description describes an example related to MPEG2TS. However, the data configuration of the multiplexed data transmitted by the transmission and reception schemes discussed in the abovedescribed Embodiments is not limited to MPEG2TS. The advantageous effects of the abovedescribed Embodiments are also achievable using any other data structure.
Each stream included in the multiplexed data is identified by an identifier, termed a PID, uniquely assigned to the stream. For example, PID 0x1011 is assigned to the video stream used for the main video of the movie, PIDs 0x1100 through 0x111F are assigned to the audio streams, PIDs 0x1200 through 0x121F are assigned to the presentation graphics, PIDs 0x1400 through 0x141F are assigned to the interactive graphics, PIDs 0x1B00 through 0x1B1F are assigned to the video streams used for the subvideo of the movie, and PIDs 0x1A00 through 0x1A1F are assigned to the audio streams used as subaudio to be mixed with the main audio of the movie.
In addition to the video streams, audio streams, presentation graphics streams, and the like, the TS packets included in the multiplexed data also include a PAT (Program Association Table), a PMT (Program Map Table), a PCR (Program Clock Reference) and so on. The PAT indicates the PID of a PMT used in the multiplexed data, and the PID of the PAT itself is registered as 0. The PMT includes PIDs identifying the respective streams, such as video, audio and subtitles, contained in the multiplexed data and attribute information (frame rate, aspect ratio, and the like) of the streams identified by the respective PIDs. In addition, the PMT includes various types of descriptors relating to the multiplexed data. One such descriptor may be copy control information indicating whether or not copying of the multiplexed data is permitted. The PCR includes information for synchronizing the ATC (Arrival Time Clock) serving as the chronological axis of the ATS to the STC (System Time Clock) serving as the chronological axis of the PTS and DTS. Each PCR packet includes an STC time corresponding to the ATS at which the packet is to be transferred to the decoder.
When recorded onto a recoding medium or the like, the multiplexed data are recorded along with a multiplexed data information file.
The multiplexed data information is made up of a system rate, a playback start time, and a playback end time. The system rate indicates the maximum transfer rate of the multiplexed data to the PID filter of a laterdescribed system target decoder. The multiplexed data includes ATS at an interval set so as not to exceed the system rate. The playback start time is set to the time specified by the PTS of the first video frame in the multiplexed data, whereas the playback end time is set to the time calculated by adding the playback duration of one frame to the PTS of the last video frame in the multiplexed data.
In the present embodiment, the stream type included in the PMT is used among the information included in the multiplexed data. When the multiplexed data are recorded on a recording medium, the video stream attribute information included in the multiplexed data information file is used. Specifically, the video coding method and device described in any of the above Embodiments may be modified to additionally include a step or unit of setting a specific piece of information in the stream type included in the PMT or in the video stream attribute information. The specific piece of information is for indicating that the video data are generated by the video coding method and device described in the Embodiment. According to such a structure, video data generated by the video coding method and device described in any of the above Embodiments is distinguishable from video data compliant with other standards.
In addition, the audiovisual output device 4500 may be operated using the Internet. For example, the audiovisual output device 4500 may be made to record (store) a program through another terminal connected to the Internet. (Accordingly, the audiovisual output device 4500 should include the drive 3708 from
The present description considers a communications/broadcasting device such as a broadcaster, a base station, an access point, a terminal, a mobile phone, or the like provided with the transmission device, and a communications device such as a television, radio, terminal, personal computer, mobile phone, access point, base station, or the like provided with the reception device. The transmission device and the reception device pertaining to the present invention are communication devices in a form able to execute applications, such as a television, radio, personal computer, mobile phone, or similar, through connection to some sort of interface (e.g., USB).
Furthermore, in the present embodiment, symbols other than data symbols, such as pilot symbols (namely preamble, unique word, postamble, reference symbols, scattered pilot symbols and so on), symbols intended for control information, and so on may be freely arranged within the frame. Although pilot symbols and symbols intended for control information are presently named, such symbols may be freely named otherwise as the function thereof remains the important consideration.
Provided that a pilot symbol, for example, is a known symbol modulated with PSK modulation in the transmitter and receiver (alternatively, the receiver may be synchronized such that the receiver knows the symbols transmitted by the transmitter), the receiver is able to use this symbol for frequency synchronization, time synchronization, channel estimation (CSI (Channel State Information) estimation for each modulated signal), signal detection, and the like.
The symbols intended for control information are symbols transmitting information (such as the modulation scheme, errorcorrecting coding scheme, coding rate of errorcorrecting codes, and setting information for the top layer used in communications) transmitted to the receiving party in order to execute transmission of nondata (i.e., applications).
The present invention is not limited to the Embodiments, but may also be realized in various other ways. For example, while the above Embodiments describe communication devices, the present invention is not limited to such devices and may be implemented as software for the corresponding communications scheme.
Although the abovedescribed Embodiments describe phase changing schemes for schemes of transmitting two modulated signals from two antennas, no limitation is intended in this regard. Precoding and a change of phase may be performed on four signals that have been mapped to generate four modulated signals transmitted using four antennas. That is, the present invention is applicable to performing a change of phase on N signals that have been mapped and precoded to generate N modulated signals transmitted using N antennas.
Although the abovedescribed Embodiments describe examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO system, the present invention is not limited in this regard and is also applicable to MISO (Multiple Input Single Output) systems. In a MISO system, the reception device does not include antenna 701_Y, wireless unit 703_Y, channel fluctuation estimator 707_1 for modulated signal z1, and channel fluctuation estimator 707_2 for modulated signal z2 from
Although the present invention describes examples of systems where two modulated signals are transmitted from two antennas and received by two respective antennas in a MIMO communications system, the present invention is not limited in this regard and is also applicable to MISO systems. In a MISO system, the transmission device performs precoding and change of phase such that the points described thus far are applicable. However, the reception device does not include antenna 701_Y, wireless unit 703_Y, channel fluctuation estimator 707_1 for modulated signal z1, and channel fluctuation estimator 707_2 for modulated signal z2 from
The present description uses terms such as precoding, precoding weights, precoding matrix, and so on. The terminology itself may be otherwise (e.g., may be alternatively termed a codebook) as the key point of the present invention is the signal processing itself.
Furthermore, although the present description discusses examples mainly using OFDM as the transmission scheme, the invention is not limited in this manner. Multicarrier schemes other than OFDM and singlecarrier schemes may all be used to achieve similar Embodiments. Here, spreadspectrum communications may also be used. When singlecarrier schemes are used, a change of phase is performed with respect to the time domain.
In addition, although the present description discusses the use of ML operations, APP, Maxlog APP, ZF, MMSE and so on by the reception device, these operations may all be generalized as wave detection, demodulation, detection, estimation, and demultiplexing as the soft results (loglikelihood and loglikelihood ratio) and the hard results (zeroes and ones) obtained thereby are the individual bits of data transmitted by the transmission device.
Different data may be transmitted by each stream s1(t) and s2(t) (s1(i), s2(i)), or identical data may be transmitted thereby.
The two stream baseband signals s1(i) and s2(i) (where i indicates sequence (with respect to time or (carrier) frequency)) undergo precoding and a regular change of phase (the order of operations may be freely reversed) to generate two postprocessing baseband signals z1(i) and z2(i). For postprocessing baseband signal z1(i), the inphase component I is I_{1}(i) while the quadrature component is Q_{1}(i), and for post processing baseband signal z2(i), the inphase component is I_{1}(i) while the quadrature component is Q_{2}(i). The baseband components may be switched, as long as the following holds.
 Let the inphase component and the quadrature component of switched baseband signal r1(i) be I_{1}(i) and Q_{2}(i), and the inphase component and the quadrature component of switched baseband signal r2(i) be I_{2}(i) and Q_{1}(i). The modulated signal corresponding to switched baseband signal r1(i) is transmitted by transmit antenna 1 and the modulated signal corresponding to switched baseband signal r2(i) is transmitted from transmit antenna 2, simultaneously on a common frequency. As such, the modulated signal corresponding to switched baseband signal r1(i) and the modulated signal corresponding to switched baseband signal r2(i) are transmitted from different antennas, simultaneously on a common frequency. Alternatively,
 For switched baseband signal r1(i), the inphase component may be I_{1}(i) while the quadrature component may be I_{2}(i), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i) while the quadrature component may be Q_{2}(i).
 For switched baseband signal r1(i), the inphase component may be I_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i) while the quadrature component may be Q_{2}(i).
 For switched baseband signal r1(i), the inphase component may be I_{1}(i) while the quadrature component may be I_{2}(i), and for switched baseband signal r2(i), the inphase component may be Q_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r1(i), the inphase component may be I_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r2(i), the inphase component may be Q_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r1(i), the inphase component may be I_{1}(i) while the quadrature component may be Q_{2}(i), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i) while the quadrature component may be I_{2}(i1).
 For switched baseband signal r1(i), the inphase component may be Q_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r2(i), the inphase component may be I_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r1(i), the inphase component may be Q_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i) while the quadrature component may be I_{2}(i).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i) while the quadrature component may be I_{2}(i), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i) while the quadrature component may be Q_{2}(i).
 For switched baseband signal r2(i), the inphase component may be I_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i) while the quadrature component may be Q_{2}(i).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i) while the quadrature component may be I_{2}(i), and for switched baseband signal r1(i), the inphase component may be Q_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r2(i), the inphase component may be I_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r1(i), the inphase component may be Q_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i) while the quadrature component may be Q_{2}(i), and for switched baseband signal r1(i), the inphase component may be I_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i) while the quadrature component may be Q_{2}(i), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i) while the quadrature component may be I_{2}(i).
 For switched baseband signal r2(i), the inphase component may be Q_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r1(i), the inphase component may be I_{2}(i) while the quadrature component may be Q_{1}(i).
 For switched baseband signal r2(i), the inphase component may be Q_{2}(i) while the quadrature component may be I_{1}(i), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i) while the quadrature component may be I_{2}(i).
Alternatively, although the above description discusses performing two types of signal processing on both stream signals so as to switch the inphase component and quadrature component of the two signals, the invention is not limited in this manner. The two types of signal processing may be performed on more than two streams, so as to switch the inphase component and quadrature component thereof.
Alternatively, although the above examples describe switching baseband signals having a common time (common (sub)carrier) frequency), the baseband signals being switched need not necessarily have a common time. For example, any of the following are possible.
 For switched baseband signal r1(i), the inphase component may be I_{1}(i+v) while the quadrature component may be Q_{2}(i+w), and for switched baseband signal r2(i), the inphase component may be I_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r1(i), the inphase component may be I_{1}(i+v) while the quadrature component may be I_{2}(i+w), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be Q_{2}(i+w).
 For switched baseband signal r1(i), the inphase component may be I_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be Q_{2}(i+w).
 For switched baseband signal r1(i), the inphase component may be I_{1}(i+v) while the quadrature component may be I_{2}(i+w), and for switched baseband signal r2(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r1(i), the inphase component may be I_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r2(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r1(i), the inphase component may be I_{1}(i+v) while the quadrature component may be Q_{2}(i+w), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be 12(i+w).
 For switched baseband signal r1(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r2(i), the inphase component may be I_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r1(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r2(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be 12(i+w).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i+v) while the quadrature component may be I_{2}(i+w), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be Q_{2}(i+w).
 For switched baseband signal r2(i), the inphase component may be I_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be Q_{2}(i+w).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i+v) while the quadrature component may be I_{2}(i+w), and for switched baseband signal r1(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r2(i), the inphase component may be I_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r1(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i+v) while the quadrature component may be Q_{2}(i+w), and for switched baseband signal r1(i), the inphase component may be I_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r2(i), the inphase component may be I_{1}(i+v) while the quadrature component may be Q_{2}(i+w), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be I_{2}(i+w).
 For switched baseband signal r2(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r1(i), the inphase component may be I_{2}(i+w) while the quadrature component may be Q_{1}(i+v).
 For switched baseband signal r2(i), the inphase component may be Q_{2}(i+w) while the quadrature component may be I_{1}(i+v), and for switched baseband signal r1(i), the inphase component may be Q_{1}(i+v) while the quadrature component may be 12(i+w).
Each of the transmit antennas of the transmission device and each of the receive antennas of the reception device shown in the figures may be formed by a plurality of antennas.
The present description uses the symbol ∀, which is the universal quantifier, and the symbol ∃, which is the existential quantifier.
Furthermore, the present description uses the radian as the unit of phase in the complex plane, e.g., for the argument thereof.
When dealing with the complex plane, the coordinates of complex numbers are expressible by way of polar coordinates. For a complex number z=a+jb (where a and b are real numbers and j is the imaginary unit), the corresponding point (a, b) on the complex plane is expressed with the polar coordinates[r, 0], converted as follows:
a=r×cos θ
b=r×sin θ
[Math. 49]
r=a^{2}+b^{2} (formula 49)
where r is the absolute value of z (r=z), and θ is the argument thereof. As such, z=a+jb is expressible as re^{jθ}.
In the present invention, the baseband signals s1, s2, z1, and z2 are described as being complex signals. A complex signal made up of inphase signal I and quadrature signal Q is also expressible as complex signal I+jQ. Here, either of I and Q may be equal to zero.
A transmitter 4607 takes the encoded video data 4602, the encoded audio data 4604, and the encoded data 4606 as input, performs errorcorrecting coding, modulation, precoding, and phase changing (e.g., the signal processing by the transmission device from
A receiver 4612 takes received signals 4611_1 through 4611_M received by antennas 4610_1 through 4610_M as input, performs processing such as frequency conversion, change of phase, decoding of the precoding, loglikelihood ratio calculation, and errorcorrecting decoding (e.g., the processing by the reception device from
In the abovedescribed Embodiments pertaining to the present invention, the number of encoders in the transmission device using a multicarrier transmission scheme such as OFDM may be any number, as described above. Therefore, as in
Although Embodiment 1 gives formula 36 as an example of a precoding matrix, another precoding matrix may also be used, when the following scheme is applied.
In the precoding matrices of formula 36 and formula 50, the value of α is set as given by formula 37 and formula 38. However, no limitation is intended in this manner. A simple precoding matrix is obtainable by setting α=1, which is also a valid value.
In Embodiment A1, the phase changers from
where k=0, 1, 2, . . . , N−2, N−1 (k denotes an integer that satisfies 0≤k≤N−1). When N=5, 7, 9, 11, or 15, the reception device is able to obtain good data reception quality.
Although the present description discusses the details of phase changing schemes involving two modulated signals transmitted by a plurality of antennas, no limitation is intended in this regard. Precoding and a change of phase may be performed on three or more baseband signals on which mapping has been performed according to a modulation scheme, followed by predetermined processing on the postphasechange baseband signals and transmission using a plurality of antennas, to realize the same results.
Programs for executing the above transmission scheme may, for example, be stored in advance in ROM (ReadOnly Memory) and be read out for operation by a CPU.
Furthermore, the programs for executing the above transmission scheme may be stored on a computerreadable recording medium, the programs stored in the recording medium may be loaded in the RAM (Random Access Memory) of the computer, and the computer may be operated in accordance with the programs.
The components of the abovedescribed Embodiments may be typically assembled as an LSI (Large Scale Integration), a type of integrated circuit. Individual components may respectively be made into discrete chips, or a subset or entirety of the components may be made into a single chip. Although an LSI is mentioned above, the terms IC (Integrated Circuit), system LSI, super LSI, or ultra LSI may also apply, depending on the degree of integration. Furthermore, the method of integrated circuit assembly is not limited to LSI. A dedicated circuit or a generalpurpose processor may be used. After LSI assembly, a FPGA (Field Programmable Gate Array) or reconfigurable processor may be used.
Furthermore, should progress in the field of semiconductors or emerging technologies lead to replacement of LSI with other integrated circuit methods, then such technology may of course be used to integrate the functional blocks. Applications to biotechnology are also plausible.
Embodiment 1 explained that the precoding matrix in use may be switched when transmission parameters change. The present embodiment describes a detailed example of such a case, where, as described above (in the supplement), the transmission parameters change such that streams s1(t) and s2(t) switch between transmitting different data and transmitting identical data, and the precoding matrix and phase changing scheme being used are switched accordingly.
The example of the present embodiment describes a situation where two modulated signals transmitted from two different transmit antenna alternate between having the modulated signals include identical data and having the modulated signals each include different data.
On the other hand, when transmitting different data, distributed data 405A are given as x1, x3, x5, x7, x9, and so on, while distributed data 405B are given as x2, x4, x6, x8, x10, and so on.
The distributor 404 determines, according to the frame configuration signal 313 taken as input, whether the transmission mode is identical data transmission or different data transmission.
An alternative to the above is shown in
One characteristic feature of the present embodiment is that, when the transmission mode switches from identical data transmission to different data transmission, the precoding matrix may also be switched. As indicated by formula 36 and formula 39 in Embodiment 1, given a matrix made up of w11, w12, w21, and w22, the precoding matrix used to transmit identical data may be as follows.
where a is a real number (a may also be a complex number, but given that the baseband signal input as a result of precoding undergoes a change of phase, a real number is preferable for considerations of circuit size and complexity reduction). Also, when a is equal to one, the weighting units 308A and 308B do not perform weighting and output the input signal asis.
Accordingly, when transmitting identical data, the weighted baseband signals 309A and 316B are identical signals output by the weighting units 308A and 308B.
When the frame configuration signal indicates identical transmission mode, a phase changer 5201 performs a change of phase on weighted baseband signal 309A and outputs postphasechange baseband signal 5202. Similarly, when the frame configuration signal indicates identical transmission mode, phase changer 317B performs a change of phase on weighted baseband signal 316B and outputs postphasechange baseband signal 309B. The change of phase performed by phase changer 5201 is of e^{jA(t) }(alternatively, e^{jA(f) }or e^{jA(t,f)}) (where t is time and f is frequency) (accordingly, e^{jA(t) }(alternatively, e^{jA(f) }or e^{jA(t,f)}) is the value by which the input baseband signal is multiplied), and the change of phase performed by phase changer 317B is of ejB(t) (alternatively, e^{jB(f) }or e^{jB(t,f)}) (where t is time and f is frequency) (accordingly, e^{jB(t) }(alternatively, e^{jB(f) }or e^{jB(t,f)}) is the value by which the input baseband signal is multiplied). As such, the following condition is satisfied.
[Math. 53]
Some time t satisfies
e^{jA(t)}≠e^{jB(t)} (formula 53)
(Or, some (carrier) frequency f satisfies e^{jA(f)}≠e^{jB(f)})
(Or, some (carrier) frequency f and time t satisfy e^{jA(t,f)}≠e^{jB(t,f)})
As such, the transmit signal is able to reduce multipath influence and thereby improve data reception quality for the reception device. (However, the change of phase may also be performed by only one of the weighted baseband signals 309A and 316B.)
In
When the selected transmission mode indicates different data transmission, then any of formula 36, formula 39, and formula 50 given in Embodiment 1 may apply. Significantly, the phase changers 5201 and 317B from
When the selected transmission mode indicates different data transmission, the precoding matrix may be as given in formula 52, or as given in any of formula 36, formula 50, and formula 39, or may be a precoding matrix unlike that given in formula 52. Thus, the reception device is especially likely to experience improvements to data reception quality in the LOS environment.
Furthermore, although the present embodiment discusses examples using OFDM as the transmission scheme, the invention is not limited in this manner. Multicarrier schemes other than OFDM and singlecarrier schemes may all be used to achieve similar Embodiments. Here, spreadspectrum communications may also be used. When singlecarrier schemes are used, the change of phase is performed with respect to the time domain.
As explained in Embodiment 3, when the transmission scheme involves different data transmission, the change of phase is performed on the data symbols, only. However, as described in the present embodiment, when the transmission scheme involves identical data transmission, then the change of phase need not be limited to the data symbols but may also be performed on pilot symbols, control symbols, and other such symbols inserted into the transmission frame of the transmit signal. (The change of phase need not always be performed on symbols such as pilot symbols and control symbols, though doing so is preferable in order to achieve diversity gain.)
The present embodiment describes a configuration scheme for a base station corresponding to Embodiment C1.
A terminal Q (5908) receives transmit signal 5903A transmitted by antenna 5904A of base station A (5902A) and transmit signal 593B transmitted by antenna 5904B of base station B (5902B), then performs predetermined processing thereon to obtained received data.
As shown, transmit signals 5903A and 5905A transmitted by base station A (5902A) and transmit signals 5903B and 5905B transmitted by base station B (5902B) use at least frequency band X and frequency band Y. Frequency band X is used to transmit data of a first channel, and frequency band Y is used to transmit data of a second channel.
Accordingly, terminal P (5907) receives transmit signal 5903A transmitted by antenna 5904A and transmit signal 5905A transmitted by antenna 5906A of base station A (5902A), extracts frequency band X therefrom, performs predetermined processing, and thus obtains the data of the first channel. Terminal Q (5908) receives transmit signal 5903A transmitted by antenna 5904A of base station A (5902A) and transmit signal 5903B transmitted by antenna 5904B of base station B (5902B), extracts frequency band Y therefrom, performs predetermined processing, and thus obtains the data of the second channel.
The following describes the configuration and operations of base station A (5902A) and base station B (5902B).
As described in Embodiment C1, both base station A (5902A) and base station B (5902B) incorporate a transmission device configured as illustrated by
The creation of encoded data in frequency band Y may involve, as shown in
Also, in
As explained above, when the base station transmits different data, the precoding matrix and phase changing scheme are set according to the transmission scheme to generate modulated signals.
On the other hand, to transmit identical data, two base stations respectively generate and transmit modulated signals. In such circumstances, base stations each generating modulated signals for transmission from a common antenna may be considered to be two combined base stations using the precoding matrix given by formula 52. The phase changing scheme is as explained in Embodiment C1, for example, and satisfies the conditions of formula 53.
In addition, the transmission scheme of frequency band X and frequency band Y may vary over time. Accordingly, as illustrated in
According to the present embodiment, not only can the reception device obtain improved data reception quality for identical data transmission as well as different data transmission, but the transmission devices can also share a phase changer.
Furthermore, although the present embodiment discusses examples using OFDM as the transmission scheme, the invention is not limited in this manner. Multicarrier schemes other than OFDM and singlecarrier schemes may all be used to achieve similar Embodiments. Here, spreadspectrum communications may also be use. When singlecarrier schemes are used, the change of phase is performed with respect to the time domain.
As explained in Embodiment 3, when the transmission scheme involves different data transmission, the change of phase is carried out on the data symbols, only. However, as described in the present embodiment, when the transmission scheme involves identical data transmission, then the change of phase need not be limited to the data symbols but may also be performed on pilot symbols, control symbols, and other such symbols inserted into the transmission frame of the transmit signal. (The change of phase need not always be performed on symbols such as pilot symbols and control symbols, though doing so is preferable in order to achieve diversity gain.)
The present embodiment describes a configuration scheme for a repeater corresponding to Embodiment C1. The repeater may also be termed a repeating station.
Repeater A (6203A) performs processing such as demodulation on received signal 6205A received by receive antenna 6204A and on received signal 6207A received by receive antenna 6206A, thus obtaining received data. Then, in order to transmit the received data to a terminal, repeater A (6203A) performs transmission processing to generate modulated signals 6209A and 6211A for transmission on respective antennas 6210A and 6212A.
Similarly, repeater B (6203B) performs processing such as demodulation on received signal 6205B received by receive antenna 6204B and on received signal 6207B received by receive antenna 6206B, thus obtaining received data. Then, in order to transmit the received data to a terminal, repeater B (6203B) performs transmission processing to generate modulated signals 6209B and 6211B for transmission on respective antennas 6210B and 6212B. Here, repeater B (6203B) is a master repeater that outputs a control signal 6208. repeater A (6203A) takes the control signal as input. A master repeater is not strictly necessary. Base station 6201 may also transmit individual control signals to repeater A (6203A) and to repeater B (6203B).
Terminal P (5907) receives modulated signals transmitted by repeater A (6203A), thereby obtaining data. Terminal Q (5908) receives signals transmitted by repeater A (6203A) and by repeater B (6203B), thereby obtaining data. Terminal R (6213) receives modulated signals transmitted by repeater B (6203B), thereby obtaining data.
As shown, the modulated signals transmitted by antenna 6202A and by antenna 6202B use at least frequency band X and frequency band Y. Frequency band X is used to transmit data of a first channel, and frequency band Y is used to transmit data of a second channel.
As described in Embodiment C1, the data of the first channel is transmitted using frequency band X in different data transmission mode. Accordingly, as shown in
As shown in
As shown in
As described in Embodiment C1, the data of the first channel is transmitted using frequency band X in different data transmission mode. Accordingly, as shown in
As shown in
The following describes the configuration of repeater A (6203A) and repeater B (6203B) from
Receiver 6203X and onward constitute a processor for generating a modulated signal for transmitting frequency band X. Further, the receiver here described is not only the receiver for frequency band X as shown in
The overall operations of the distributor 404 are identical to those of the distributor in the base station described in Embodiment C2.
When transmitting as indicated in
As for frequency band Y, repeater A (6203A) operates a processor 6500 pertaining to frequency band Y and corresponding to the signal processor 6500 pertaining to frequency band X shown in
As shown in
As explained above, when the repeater transmits different data, the precoding matrix and phase changing scheme are set according to the transmission scheme to generate modulated signals.
On the other hand, to transmit identical data, two repeaters respectively generate and transmit modulated signals. In such circumstances, repeaters each generating modulated signals for transmission from a common antenna may be considered to be two combined repeaters using the precoding matrix given by formula 52. The phase changing scheme is as explained in Embodiment C1, for example, and satisfies the conditions of formula 53.
Also, as explained in Embodiment C1 for frequency band X, the base station and repeater may each have two antennas that transmit respective modulated signals and two antennas that receive identical data. The operations of such a base station or repeater are as described for Embodiment C1.
According to the present embodiment, not only can the reception device obtain improved data reception quality for identical data transmission as well as different data transmission, but the transmission devices can also share a phase changer.
Furthermore, although the present embodiment discusses examples using OFDM as the transmission scheme, the invention is not limited in this manner. Multicarrier schemes other than OFDM and singlecarrier schemes may all be used to achieve similar Embodiments. Here, spreadspectrum communications may also be used. When singlecarrier schemes are used, the change of phase is performed with respect to the time domain.
As explained in Embodiment 3, when the transmission scheme involves different data transmission, the change of phase is carried out on the data symbols, only. However, as described in the present embodiment, when the transmission scheme involves identical data transmission, then the change of phase need not be limited to the data symbols but may also be performed on pilot symbols, control symbols, and other such symbols inserted into the transmission frame of the transmit signal. (The change of phase need not always be performed on symbols such as pilot symbols and control symbols, though doing so is preferable in order to achieve diversity gain.)
The present embodiment concerns a phase changing scheme different from the phase changing schemes described in Embodiment 1 and in the Supplement.
In Embodiment 1, formula 36 is given as an example of a precoding matrix, and in the Supplement, formula 50 is similarly given as another such example. In Embodiment A1, the phase changers from
where k=0, 1, 2, . . . , N−2, N−1 (k denotes an integer that satisfies 0≤k≤N−1).
Accordingly, the reception device is able to achieve improvements in data reception quality in the LOS environment, and especially in a radio wave propagation environment. In the LOS environment, when the change of phase has not been performed, a regular phase relationship holds. However, when the change of phase is performed, the phase relationship is modified, in turn avoiding poor conditions in a burstlike propagation environment. As an alternative to formula 54, PHASE[k] may be calculated as follows.
where k=0, 1, 2, . . . , N−2, N−1 (k denotes an integer that satisfies 0≤k≤N−1).
As a further alternative phase changing scheme, PHASE[k] may be calculated as follows.
where k=0, 1, 2, . . . , N−2, N−1 (k denotes an integer that satisfies 0≤k≤N−1), and Z is a fixed value.
As a further alternative phase changing scheme, PHASE[k] may be calculated as follows.
where k=0, 1, 2, . . . , N−2, N−1 (k denotes an integer that satisfies 0≤k≤N−1), and Z is a fixed value.
As such, by performing the change of phase according to the present embodiment, the reception device is made more likely to obtain good reception quality.
The change of phase of the present embodiment is applicable not only to singlecarrier schemes but also to multicarrier schemes. Accordingly, the present embodiment may also be realized using, for example, spreadspectrum communications, OFDM, SCFDMA, SCOFDM, wavelet OFDM as described in NonPatent Literature 7, and so on. As previously described, while the present embodiment explains the change of phase by changing the phase with respect to the time domain t, the phase may alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the change of phase in the time domain t described in the present embodiment and replacing t with f (f being the ((sub) carrier) frequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing scheme of the present embodiment is also applicable to a change of phase in both the time domain and the frequency domain. Further, when the phase changing scheme described in the present embodiment satisfies the conditions indicated in Embodiment A1, the reception device is highly likely to obtain good data quality.
The present embodiment concerns a phase changing scheme different from the phase changing schemes described in Embodiment 1, in the Supplement, and in Embodiment C4.
In Embodiment 1, formula 36 is given as an example of a precoding matrix, and in the Supplement, formula 50 is similarly given as another such example. In Embodiment A1, the phase changers from
The characteristic feature of the phase changing scheme pertaining to the present embodiment is the period (cycle) of N=2n+1. To achieve the period (cycle) of N=2n+1, n+1 different phase changing values are prepared. Among these n+1 different phase changing values, n phase changing values are used twice per period (cycle), and one phase changing value is used only once per period (cycle), thus achieving the period (cycle) of N=2n+1. The following describes these phase changing values in detail.
The n+1 different phase changing values required to achieve a phase changing scheme in which the phase changing value is regularly switched in a period (cycle) of N=2n+1 are expressed as PHASE[0], PHASE[1], PHASE[i], . . . , PHASE[n−1], PHASE[n] (where i=0, 1, 2, . . . , n−2, n−1, n (i denotes an integer that satisfies 0≤i≤n)). Here, the n+1 different phase changing values of PHASE[0], PHASE[1], PHASE[i], . . . , PHASE[n−1], PHASE[n] are expressed as follows.
where k=0, 1, 2, . . . , n−2, n−1, n (k denotes an integer that satisfies 0≤k≤n). The n+1 different phase changing values PHASE[0], PHASE[1], . . . , PHASE[i], . . . , PHASE[n−1], PHASE[n] are given by formula 58. PHASE[0] is used once, while PHASE[1] through PHASE[n] are each used twice (i.e., PHASE[1] is used twice, PHASE[2] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing scheme in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing scheme is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are fewer, the effect thereof on the transmission device and reception device may be reduced. According to the above, the reception device is able to achieve improvements in data reception quality in the LOS environment, and especially in a radio wave propagation environment. In the LOS environment, when the change of phase has not been performed, a regular phase relationship occurs. However, when the change of phase is performed, the phase relationship is modified, in turn avoiding poor conditions in a burstlike propagation environment. As an alternative to formula 54, PHASE[k] may be calculated as follows.
where k=0, 1, 2, . . . , n−2, n−1, n (k denotes an integer that satisfies 0≤k≤n).
The n+1 different phase changing values PHASE[0], PHASE[1], PHASE[i], . . . , PHASE[n−1], PHASE[n] are given by formula 59. PHASE[0] is used once, while PHASE[1] through PHASE[n] are each used twice (i.e., PHASE[1] is used twice, PHASE[2] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing scheme in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing scheme is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are fewer, the effect thereof on the transmission device and reception device may be reduced.
As a further alternative, PHASE[k] may be calculated as follows.
where k=0, 1, 2, . . . , n−2, n−1, n (k denotes an integer that satisfies 0≤k≤n) and Z is a fixed value.
The n+1 different phase changing values PHASE[0], PHASE[1], PHASE[i], . . . , PHASE[n−1], PHASE[n] are given by formula 60. PHASE[0] is used once, while PHASE[1] through PHASE[n] are each used twice (i.e., PHASE[1] is used twice, PHASE[2] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing scheme in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing scheme is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are fewer, the effect thereof on the transmission device and reception device may be reduced.
As a further alternative, PHASE[k] may be calculated as follows.
where k=0, 1, 2, . . . , n−2, n−1, n(k denotes an integer that satisfies 0≤k≤n) and Z is a fixed value.
The n+1 different phase changing values PHASE[0], PHASE[1], PHASE[i], . . . , PHASE[n−1], PHASE[n] are given by formula 61. PHASE[0] is used once, while PHASE[1] through PHASE[n] are each used twice (i.e., PHASE[1] is used twice, PHASE[2] is used twice, and so on, until PHASE[n−1] is used twice and PHASE[n] is used twice). As such, through this phase changing scheme in which the phase changing value is regularly switched in a period (cycle) of N=2n+1, a phase changing scheme is realized in which the phase changing value is regularly switched between fewer phase changing values. Thus, the reception device is able to achieve better data reception quality. As the phase changing values are smaller, the effect thereof on the transmission device and reception device may be reduced.
As such, by performing the change of phase according to the present embodiment, the reception device is made more likely to obtain good reception quality.
The change of phase of the present embodiment is applicable not only to singlecarrier schemes but also to transmission using multicarrier schemes. Accordingly, the present embodiment may also be realized using, for example, spreadspectrum communications, OFDM, SCFDMA, SCOFDM, wavelet OFDM as described in NonPatent Literature 7, and so on. As previously described, while the present embodiment explains the change of phase as a change of phase with respect to the time domain t, the phase may alternatively be changed with respect to the frequency domain as described in Embodiment 1. That is, considering the change of phase with respect to the time domain t described in the present embodiment and replacing t with f (f being the ((sub) carrier) frequency) leads to a change of phase applicable to the frequency domain. Also, as explained above for Embodiment 1, the phase changing scheme of the present embodiment is also applicable to a change of phase with respect to both the time domain and the frequency domain.
The present embodiment describes a scheme for regularly changing the phase, specifically that of Embodiment C5, when encoding is performed using block codes as described in NonPatent Literature 12 through 15, such as QC LDPC Codes (not only QCLDPC but also LDPC codes may be used), concatenated LDPC (blocks) and BCH codes, Turbo codes or DuoBinary Turbo Codes using tailbiting, and so on. The following example considers a case where two streams s1 and s2 are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information.
As shown in
Then, given that the transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 750 slots are needed to transmit all of the bits making up one coded block, and when the modulation scheme is 64QAM, 500 slots are needed to transmit all of the bits making up one coded block.
The following describes the relationship between the abovedefined slots and the phase, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase, which has a period (cycle) of five. That is, the phase changer of the transmission device from
The following describes the relationship between the abovedefined slots and the phase, as pertains to schemes for a regular change of phase.
For the abovedescribed 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, phase changing value P[0] is used on 300 slots, phase changing value P[1] is used on 300 slots, phase changing value P[2] is used on 300 slots, phase changing value P[3] is used on 300 slots, and phase changing value P[4] is used on 300 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality.
Similarly, for the abovedescribed 750 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16QAM, phase changing value P[0] is used on 150 slots, phase changing value P[1] is used on 150 slots, phase changing value P[2] is used on 150 slots, phase changing value P[3] is used on 150 slots, and phase changing value P[4] is used on 150 slots.
Furthermore, for the abovedescribed 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64QAM, phase changing value P[0] is used on 100 slots, phase changing value P[1] is used on 100 slots, phase changing value P[2] is used on 100 slots, phase changing value P[3] is used on 100 slots, and phase changing value P[4] is used on 100 slots.
As described above, a phase changing scheme for a regular change of phase changing value as given in Embodiment C5 requires the preparation of N=2n+1 phase changing values P[0], P[1], . . . , P[2n−1], P[2n] (where P[0], P[1], . . . , P[2n−1], P[2n] are expressed as PHASE[0], PHASE[1], PHASE[2], . . . , PHASE[n−1], PHASE[n] (see Embodiment C5)). As such, in order to transmit all of the bits making up a single coded block, phase changing value P[0] is used on K_{0 }slots, phase changing value P[1] is used on K_{1 }slots, phase changing value P[i] is used on K_{i }slots (where i=0, 1, 2, . . . , 2n−1, 2n (i denotes an integer that satisfies 0≤i≤2n)), and phase changing value P[2n] is used on K_{2}n slots, such that Condition #C01 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K_{2n}. That is, K_{a}=K_{b }(∀a and ∀b where a, b, =0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b).
A phase changing scheme for a regular change of phase changing value as given in Embodiment C5 having a period (cycle) of N=2n+1 requires the preparation of phase changing values PHASE[0], PHASE[1], PHASE[2], PHASE[n−1], PHASE[n]. As such, in order to transmit all of the bits making up a single coded block, phase changing value PHASE[0] is used on G_{0 }slots, phase changing value PHASE[1] is used on G_{1 }slots, phase changing value PHASE[i] is used on Gi slots (where i=0, 1, 2, . . . , n−1, n (i denotes an integer that satisfies 0≤i≤n), and phase changing value PHASE[n] is used on G_{n }slots, such that Condition #C01 is met. Condition #C01 may be modified as follows.
2×G_{0}=G_{1 }. . . =G_{1}= . . . G_{n}. That is, 2×G_{0}=G_{a }(∀a where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
Then, when a communication system that supports multiple modulation schemes selects one such supported scheme for use, Condition #C01 (or Condition #C02) should preferably be met for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C01 (or Condition #C02) may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #C01.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n) a≠b). Alternatively, Condition #C03 may be expressed as follows.
The difference between G_{a }and G_{b }satisfies 0, 1, or 2. That is, G_{a}−G_{b} satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes an integer that satisfies 1≤b≤n), a≠b)
and
The difference between 2×G_{0 }and G_{a }satisfies 0, 1, or 2. That is, 2×G_{0}−G_{a} satisfies 0, 1, or 2 (∀a, where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
As shown in
The transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 1500 slots are needed to transmit all of the bits making up one coded block, and when the modulation scheme is 64QAM, 1000 slots are needed to transmit all of the bits making up one coded block.
The following describes the relationship between the abovedefined slots and the phase, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase, which has a period (cycle) of five. That is, the phase changer of the transmission device from
For the abovedescribed 3000 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is QPSK, phase changing value P[0] is used on 600 slots, phase changing value P[1] is used on 600 slots, phase changing value P[2] is used on 600 slots, phase changing value P[3] is used on 600 slots, and phase changing value P[4] is used on 600 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality.
Further, in order to transmit the first coded block, phase changing value P[0] is used on slots 600 times, phase changing value P[1] is used on slots 600 times, phase changing value P[2] is used on slots 600 times, phase changing value P[3] is used on slots 600 times, and phase changing value PHASE[4] is used on slots 600 times. Furthermore, in order to transmit the second coded block, phase changing value P[0] is used on slots 600 times, phase changing value P[1] is used on slots 600 times, phase changing value P[2] is used on slots 600 times, phase changing value P[3] is used on slots 600 times, and phase changing value P[4] is used on slots 600 times.
Similarly, for the abovedescribed 1500 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is 16QAM, phase changing value P[0] is used on 300 slots, phase changing value P[1] is used on 300 slots, phase changing value P[2] is used on 300 slots, phase changing value P[3] is used on 300 slots, and phase changing value P[4] is used on 300 slots.
Furthermore, in order to transmit the first coded block, phase changing value P[0] is used on slots 300 times, phase changing value P[1] is used on slots 300 times, phase changing value P[2] is used on slots 300 times, phase changing value P[3] is used on slots 300 times, and phase changing value P[4] is used on slots 300 times. Furthermore, in order to transmit the second coded block, phase changing value P[0] is used on slots 300 times, phase changing value P[1] is used on slots 300 times, phase changing value P[2] is used on slots 300 times, phase changing value P[3] is used on slots 300 times, and phase changing value P[4] is used on slots 300 times.
Furthermore, for the abovedescribed 1000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 64QAM, phase changing value P[0] is used on 200 slots, phase changing value P[1]is used on 200 slots, phase changing value P[2] is used on 200 slots, phase changing value P[3] is used on 200 slots, and phase changing value P[4] is used on 200 slots.
Further, in order to transmit the first coded block, phase changing value P[0] is used on slots 200 times, phase changing value P[1] is used on slots 200 times, phase changing value P[2] is used on slots 200 times, phase changing value P[3] is used on slots 200 times, and phase changing value P[4] is used on slots 200 times. Furthermore, in order to transmit the second coded block, phase changing value P[0] is used on slots 200 times, phase changing value P[1] is used on slots 200 times, phase changing value P[2] is used on slots 200 times, phase changing value P[3] is used on slots 200 times, and phase changing value P[4] is used on slots 200 times.
As described above, a phase changing scheme for regularly varying the phase changing value as given in Embodiment C5 requires the preparation of N=2n+1 phase changing values P[0], P[1], . . . , P[2n1], P[2n] (where P[0], P[1], . . . , P[2n−1], P[2n] are expressed as PHASE[0], PHASE[1], PHASE[2], PHASE[n−1], PHASE[n] (see Embodiment C5)). As such, in order to transmit all of the bits making up the two coded blocks, phase changing value P[0] is used on K_{0 }slots, phase changing value P[1] is used on K_{1 }slots, phase changing value P[i] is used on K_{i }slots (where i=0, 1, 2, . . . , 2n−1, 2n (i denotes an integer that satisfies 0≤i≤2n)), and phase changing value P[2n] is used on K_{2}n slots, such that Condition #C01 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K_{2n}. That is, K_{a}=K_{b }(∀a and ∀b where a, b, =0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a b). In order to transmit all of the bits making up the first coded block, phase changing value P[0] is used K_{0,1 }times, phase changing value P[1] is used K_{1,1 }times, phase changing value P[i] is used K_{i,1 }(where i=0, 1, 2, . . . , 2n−1, 2n (i denotes an integer that satisfies 0≤i≤2n)), and phase changing value P[2n] is used K_{2n,1 }times.
K_{0,1}=K_{1,1 }. . . =K_{i,1}= . . . K_{2n,1}. That is, K_{a,1}=K_{b,1 }(∀a and ∀b where a, b, =0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b).
In order to transmit all of the bits making up the second coded block, phase changing value P[0] is used K_{0,2 }times, phase changing value P[1] is used K_{1,2 }times, phase changing value P[i] is used K_{i,2 }(where i=0, 1, 2, . . . , 2n−1, 2n (i denotes an integer that satisfies 0≤i≤2n)), and phase changing value P[2n] is used K_{2n,2 }times.
K_{0,2}=K_{1,2 }. . . =K_{i,2}= . . . K_{2n,2}. That is, K_{a,2}=K_{b,2 }(∀a and ∀b where a, b, =0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b).
A phase changing scheme for regularly varying the phase changing value as given in Embodiment C5 having a period (cycle) of N=2n+1 requires the preparation of phase changing values PHASE[0], PHASE[1], PHASE[2], PHASE[n−1], PHASE[n]. As such, in order to transmit all of the bits making up the two coded blocks, phase changing value PHASE[0] is used on G_{0 }slots, phase changing value PHASE[1] is used on G_{1 }slots, phase changing value PHASE[i] is used on Gi slots (where i=0, 1, 2, . . . , n−1, n (i denotes an integer that satisfies 0≤i≤n)), and phase changing value PHASE[n] is used on G_{n }slots, such that Condition #C05 is met.
2×G_{0}=G_{1 }. . . =G_{i}= . . . G_{n}. That is, 2×G_{0}=G_{a }(∀a where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes an integer that satisfies 1≤b≤n)).
In order to transmit all of the bits making up the first coded block, phase changing value PHASE[0] is used G_{0,1 }times, phase changing value PHASE[1] is used G_{1,1 }times, phase changing value PHASE[i] is used G_{i,1 }(where i=0, 1, 2, . . . , n−1, n (i denotes an integer that satisfies 0≤i≤n)), and phase changing value PHASE[n] is used G_{n,1 }times.
2×G_{0,1}=G_{1,1 }. . . =G_{i,1}=G_{n,1}. That is, 2×G_{0,1}=G_{a,1 }(∀a where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
In order to transmit all of the bits making up the second coded block, phase changing value PHASE[0] is used G_{0,2 }times, phase changing value PHASE[1] is used G_{1,2 }times, phase changing value PHASE[i] is used G_{i,2 }(where i=0, 1, 2, . . . , n−1, n (i denotes an integer that satisfies 0≤i≤n)), and phase changing value PHASE[n] is used G_{n,1 }times.
2×G_{0,2}=G_{1,2 }. . . =G_{i,2 }. . . G_{n,2}. That is, 2×G_{0,2}=G_{a,2 }(∀a where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
Then, when a communication system that supports multiple modulation schemes selects one such supported scheme for use, Condition #C05, Condition #C06, and Condition #C07 (or Condition #C08, Condition #C09, and Condition #C10) should preferably be met for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C05, Condition #C06, and Condition #C07 (or Condition #C08, Condition #C09, and Condition #C10) may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #C05, Condition #C06, and Condition #C07.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b).
The difference between K_{a,1 }and K_{b,1 }satisfies 0 or 1. That is, K_{a,1}−K_{b,1} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b).
The difference between K_{a,2 }and K_{b,2 }satisfies 0 or 1. That is, K_{a,2}−K_{b,2} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , 2n−1, 2n (a denotes an integer that satisfies 0≤a≤2n, b denotes an integer that satisfies 0≤b≤2n), a≠b). Alternatively, Condition #C11, Condition #C12, and Condition #C13 may be expressed as follows.
The difference between G_{a }and G_{b }satisfies 0, 1, or 2. That is, G_{a}G_{b} satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes an integer that satisfies 1≤b≤n), a≠b)
and
The difference between 2×G_{0 }and G_{a }satisfies 0, 1, or 2. That is, 2×G_{0}G_{a} satisfies 0, 1, or 2 (∀a, where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
The difference between G_{a,1 }and G_{b},i satisfies 0, 1, or 2. That is, G_{a,1}−G_{b,1} satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes an integer that satisfies 1≤b≤n), a≠b)
and
The difference between 2×G_{0,1 }and G_{a,1 }satisfies 0, 1, or 2. That is, 2×G_{0,1}−G_{a,1} satisfies 0, 1, or 2 (∀a, where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
The difference between G_{a,2 }and G_{b,2 }satisfies 0, 1, or 2. That is, G_{a,2}−G_{b,2} satisfies 0, 1, or 2 (∀a, ∀b, where a, b=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n, b denotes an integer that satisfies 1≤b≤n), a≠b)
and
The difference between 2×G_{0,2 }and G_{a,2 }satisfies 0, 1, or 2. That is, 2×G_{0,2}−G_{a,2} satisfies 0, 1, or 2 (∀a, where a=1, 2, . . . , n−1, n (a denotes an integer that satisfies 1≤a≤n)).
As described above, bias among the phase changing values being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase changing values. As such, data reception quality can be improved for the reception device.
In the present embodiment, N phase changing values (or phase changing sets) are needed in order to perform the change of phase having a period (cycle) of N with a regular phase changing scheme. As such, N phase changing values (or phase changing sets) P[0], P[1], P[2], . . . , P[N−2], and P[N−1] are prepared. However, schemes exist for ordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) P[0], P[1], P[2], . . . , P[N−2], and P[N−1] may also change the phases of blocks in the time domain or in the timefrequency domain to obtain a symbol arrangement as described in Embodiment 1. Although the above examples discuss a phase changing scheme with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the abovedescribed conditions are satisfied, quality data reception improvements are realizable for the reception device.
Furthermore, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase, the transmission device (broadcaster, base station) may select any one of these transmission schemes.
As described in NonPatent Literature 3, spatial multiplexing MIMO schemes involve transmitting signals s1 and s2, which are mapped using a selected modulation scheme, on each of two different antennas. MIMO schemes using a fixed precoding matrix involve performing precoding only (with no change of phase). Further, spacetime block coding schemes are described in NonPatent Literature 9, 16, and 17. Singlestream transmission schemes involve transmitting signal s1, mapped with a selected modulation scheme, from an antenna after performing predetermined processing.
Schemes using multicarrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multicarrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub)carrier group are preferably used to realize the present embodiment.
When a change of phase by, for example, a phase changing value for P[i] of X radians is performed on only one precoded baseband signal, the phase changers from
The present embodiment describes a scheme for regularly changing the phase, specifically as done in Embodiment A1 and Embodiment C6, when encoding is performed using block codes as described in NonPatent Literature 12 through 15, such as QC LDPC Codes (not only QCLDPC but also LDPC (block) codes may be used), concatenated LDPC and BCH codes, Turbo codes or DuoBinary Turbo Codes, and so on. The following example considers a case where two streams s_{1 }and s2 are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information.
As shown in
Then, given that the transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 750 slots are needed to transmit all of the bits making up one coded block, and when the modulation scheme is 64QAM, 500 slots are needed to transmit all of the bits making up one coded block.
The following describes the relationship between the abovedefined slots and the phase, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase, which has a period (cycle) of five. The phase changing values (or phase changing sets) prepared in order to regularly change the phase with a period (cycle) of five are P[0], P[1], P[2], P[3], and P[4]. However, P[0], P[1], P[2], P[3], and P[4] should include at least two different phase changing values (i.e., P[0], P[1], P[2], P[3], and P[4] may include identical phase changing values). (As in
For the abovedescribed 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, phase changing value P[0] is used on 300 slots, phase changing value P[1] is used on 300 slots, phase changing value P[2] is used on 300 slots, phase changing value P[3] is used on 300 slots, and phase changing value P[4] is used on 300 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality.
Furthermore, for the abovedescribed 750 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16QAM, phase changing value P[0] is used on 150 slots, phase changing value P[1] is used on 150 slots, phase changing value P[2] is used on 150 slots, phase changing value P[3] is used on 150 slots, and phase changing value P[4] is used on 150 slots.
Further, for the abovedescribed 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64QAM, phase changing value P[0] is used on 100 slots, phase changing value P[1] is used on 100 slots, phase changing value P[2] is used on 100 slots, phase changing value P[3] is used on 100 slots, and phase changing value P[4] is used on 100 slots.
As described above, the phase changing values used in the phase changing scheme regularly switching between phase changing values with a period (cycle) of N are expressed as P[0], P[1], . . . , P[N−2], P[N−1]. However, P[0], P[1], . . . , P[N−2], P[N−1] should include at least two different phase changing values (i.e., P[0], P[1], . . . , P[N−2], P[N−1] may include identical phase changing values). In order to transmit all of the bits making up a single coded block, phase changing value P[0] is used on K_{0 }slots, phase changing value P[1] is used on K_{1 }slots, phase changing value P[i] is used on K_{i }slots (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and phase changing value P[N−1] is used on K_{N−1 }slots, such that Condition #C17 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K_{N−1}. That is, K_{a}=K_{b }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Then, when a communication system that supports multiple modulation schemes selects one such supported scheme for use, Condition #C17 should preferably be met for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C17 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #C17.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
As shown in
The transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 1500 slots are needed to transmit all of the bits making up one coded block, and when the modulation scheme is 64QAM, 1000 slots are needed to transmit all of the bits making up one coded block.
The following describes the relationship between the abovedefined slots and the phase, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase, which has a period (cycle) of five. That is, the phase changer of the transmission device from
For the abovedescribed 3000 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is QPSK, phase changing value P[0] is used on 600 slots, phase changing value P[1] is used on 600 slots, phase changing value P[2] is used on 600 slots, phase changing value P[3] is used on 600 slots, and phase changing value P[4] is used on 600 slots. This is due to the fact that any bias in phase changing value usage causes great influence to be exerted by the more frequently used phase changing value, and that the reception device is dependent on such influence for data reception quality.
Further, in order to transmit the first coded block, phase changing value P[0] is used on slots 600 times, phase changing value P[1] is used on slots 600 times, phase changing value P[2] is used on slots 600 times, phase changing value P[3] is used on slots 600 times, and phase changing value P[4] is used on slots 600 times. Furthermore, in order to transmit the second coded block, phase changing value P[0] is used on slots 600 times, phase changing value P[1] is used on slots 600 times, phase changing value P[2] is used on slots 600 times, phase changing value P[3] is used on slots 600 times, and phase changing value P[4] is used on slots 600 times.
Similarly, for the abovedescribed 1500 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is 16QAM, phase changing value P[0] is used on 300 slots, phase changing value P[1] is used on 300 slots, phase changing value P[2] is used on 300 slots, phase changing value P[3] is used on 300 slots, and phase changing value P[4] is used on 300 slots.
Further, in order to transmit the first coded block, phase changing value P[0] is used on slots 300 times, phase changing value P[1] is used on slots 300 times, phase changing value P[2] is used on slots 300 times, phase changing value P[3] is used on slots 300 times, and phase changing value P[4] is used on slots 300 times. Furthermore, in order to transmit the second coded block, phase changing value P[0] is used on slots 300 times, phase changing value P[1] is used on slots 300 times, phase changing value P[2] is used on slots 300 times, phase changing value P[3] is used on slots 300 times, and phase changing value P[4] is used on slots 300 times.
Similarly, for the abovedescribed 1000 slots needed to transmit the 6000×2 bits making up the pair of coded blocks when the modulation scheme is 64QAM, phase changing value P[0] is used on 200 slots, phase changing value P[1] is used on 200 slots, phase changing value P[2] is used on 200 slots, phase changing value P[3] is used on 200 slots, and phase changing value P[4] is used on 200 slots.
Further, in order to transmit the first coded block, phase changing value P[0] is used on slots 200 times, phase changing value P[1] is used on slots 200 times, phase changing value P[2] is used on slots 200 times, phase changing value P[3] is used on slots 200 times, and phase changing value P[4] is used on slots 200 times. Furthermore, in order to transmit the second coded block, phase changing value P[0] is used on slots 200 times, phase changing value P[1] is used on slots 200 times, phase changing value P[2] is used on slots 200 times, phase changing value P[3] is used on slots 200 times, and phase changing value P[4] is used on slots 200 times.
As described above, the phase changing values used in the phase changing scheme regularly switching between phase changing values with a period (cycle) of N are expressed as P[0], P[1], . . . , P[N−2], P[N−1]. However, P[0], P[1], . . . , P[N−2], P[N−1] should include at least two different phase changing values (i.e., P[0], P[1], . . . , P[N−2], P[N−1] may include identical phase changing values). In order to transmit all of the bits making up two coded blocks, phase changing value P[0] is used on K_{0 }slots, phase changing value P[1] is used on K_{1 }slots, phase changing value P[i] is used on K_{i }slots (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and phase changing value P[N−1] is used on K_{N−1 }slots, such that Condition #C19 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K_{N−1}. That is, K_{a}=K_{b }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
In order to transmit all of the bits making up the first coded block, phase changing value P[0] is used K_{0,1 }times, phase changing value P[1] is used K_{i},1 times, phase changing value P[i] is used K_{i,1 }(where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and phase changing value P[N−1] is used K_{N−1,1 }times.
K_{0,1}=K_{1,1}= . . . K_{i,1}= . . . K_{N−1,1}. That is, K_{a,1}=K_{b,1 }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
In order to transmit all of the bits making up the second coded block, phase changing value P[0] is used K_{0,2 }times, phase changing value P[1] is used K_{1,2 }times, phase changing value P[i] is used K_{i,2 }(where i=0, 1, 2, . . . , N−1(i denotes an integer that satisfies 0≤i≤N−1)), and phase changing value P[N−1] is used K_{N−1,2 }times.
K_{0,2}=K_{1,2}= . . . K_{i,2}= . . . K_{N−1,2}. That is, K_{a,2}=K_{b,2 }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Then, when a communication system that supports multiple modulation schemes selects one such supported scheme for use, Condition #C19, Condition #C20, and Condition #C21 are preferably met for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #C19, Condition #C20, and Condition #C21 may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #C19, Condition #C20, and Condition #C21.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
The difference between K_{a,1 }and K_{b,1 }satisfies 0 or 1. That is, K_{a}−K_{b} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
The difference between K_{a,2 }and K_{b,2 }satisfies 0 or 1. That is, K_{a,2}−K_{b,2} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
As described above, bias among the phase changing values being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase changing values. As such, data reception quality can be improved for the reception device.
In the present embodiment, N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the scheme for a regular change of phase. As such, N phase changing values (or phase changing sets) P[0], P[1], P[2], . . . , P[N−2], and P[N−1] are prepared. However, schemes exist for ordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) P[0], P[1], P[2], . . . , P[N−2], and P[N−1] may also change the phases of blocks in the time domain or in the timefrequency domain to obtain a symbol arrangement as described in Embodiment 1. Although the above examples discuss a phase changing scheme with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the abovedescribed conditions are satisfied, great quality data reception improvements are realizable for the reception device.
Furthermore, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase, the transmission device (broadcaster, base station) may select any one of these transmission schemes.
As described in NonPatent Literature 3, spatial multiplexing MIMO schemes involve transmitting signals s1 and s2, which are mapped using a selected modulation scheme, on each of two different antennas. MIMO schemes using a fixed precoding matrix involve performing precoding only (with no change of phase). Further, spacetime block coding schemes are described in NonPatent Literature 9, 16, and 17. Singlestream transmission schemes involve transmitting signal s1, mapped with a selected modulation scheme, from an antenna after performing predetermined processing.
Schemes using multicarrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multicarrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub)carrier group are preferably used to realize the present embodiment.
When a change of phase by, for example, a phase changing value for P[i] of X radians is performed on only one precoded baseband signal, the phase changers of
The present embodiment is first described as a variation of Embodiment 1.
Here, the precoding matrix is
In formula 62 above,
α is given by formula 63.
Alternatively, in formula 62,
α may be given by formula 64.
Alternatively, the precoding matrix is not restricted to that of formula 62, but may also be:
where a=Ae^{iδ11}, b=Be^{jδ12}, c=Ce^{jδ21}, and d=De^{jδ22}. Further, one of a, b, c, and d may be equal to zero. For example: (1) a may be zero while b, c, and d are nonzero, (2) b may be zero while a, c, and d are nonzero, (3) c may be zero while a, b, and d are nonzero, or (4) d may be zero while a, b, and c are nonzero.
Alternatively, any two of a, b, c, and d may be equal to zero. For example, (1) a and d may be zero while b and c are nonzero, or (2) b and c may be zero while a and d are nonzero.
When any of the modulation scheme, errorcorrecting codes, and the coding rate thereof are changed, the precoding matrix in use may also be set and changed, or the same precoding matrix may be used asis.
Next, the baseband signal switcher 6702 from
In
Here, the baseband components are switched by the baseband signal switcher 6702, such that:
 For switched baseband signal q1(i), the inphase component I may be I_{p1}(i) while the quadrature component Q may be Q_{p2}(i), and for switched baseband signal q2(i), the inphase component I may be I_{p2}(i) while the quadrature component q may be Q_{p1}(i). The modulated signal corresponding to switched baseband signal q1(i) is transmitted by transmit antenna 1 and the modulated signal corresponding to switched baseband signal q2(i) is transmitted from transmit antenna 2, simultaneously on a common frequency. As such, the modulated signal corresponding to switched baseband signal q1(i) and the modulated signal corresponding to switched baseband signal q2(i) are transmitted from different antennas, simultaneously on a common frequency. Alternatively,
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i) while the quadrature component may be I_{p2}(i), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i) while the quadrature component may be Qp_{2}(i).
 For switched baseband signal q1(i), the inphase component may be Ip2(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i) while the quadrature component may be Q_{p2}(i).
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i) while the quadrature component may be I_{p2}(i), and for switched baseband signal q2(i), the inphase component may be Q_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q1(i), the inphase component may be I_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q2(i), the inphase component may be Q_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i) while the quadrature component may be Q_{p2}(i), and for switched baseband signal q_{2}(i), the inphase component may be Q_{p1}(i) while the quadrature component may be Ip_{2}(i).
 For switched baseband signal q1(i), the inphase component may be Q_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q2(i), the inphase component may be I_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q1(i), the inphase component may be Q_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i) while the quadrature component may be I_{p2}(i).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i) while the quadrature component may be I_{p2}(i), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i) while the quadrature component may be Qp2(i).
 For switched baseband signal q2(i), the inphase component may be I_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i) while the quadrature component may be Q_{p2}(i).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i) while the quadrature component may be I_{p2}(i), and for switched baseband signal q1(i), the inphase component may be Q_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q2(i), the inphase component may be I_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q1(i), the inphase component may be Q_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i) while the quadrature component may be Q_{p2}(i), and for switched baseband signal q1(i), the inphase component may be I_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i) while the quadrature component may be Q_{p2}(i), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i) while the quadrature component may be I_{p2}(i).
 For switched baseband signal q2(i), the inphase component may be Q_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q1(i), the inphase component may be I_{p2}(i) while the quadrature component may be Q_{p1}(i).
 For switched baseband signal q2(i), the inphase component may be Q_{p2}(i) while the quadrature component may be I_{p1}(i), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i) while the quadrature component may be I_{p2}(i).
Alternatively, the weighted signals 309A and 316B are not limited to the abovedescribed switching of inphase component and quadrature component. Switching may be performed on inphase components and quadrature components greater than those of the two signals.
Also, while the above examples describe switching performed on baseband signals having a common time (common (sub)carrier) frequency), the baseband signals being switched need not necessarily have a common time (common (sub)carrier) frequency). For example, any of the following are possible.
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w), and for switched baseband signal q2(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be I_{p2}(i+w), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w).
 For switched baseband signal q1(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w).
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be I_{p2}(i+w), and for switched baseband signal q2(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q1(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q2(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q1(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be I_{p2}(i+w).
 For switched baseband signal q1(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q2(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q1(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q2(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be I_{p2}(i+w).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be I_{p2}(i+w), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w).
 For switched baseband signal q2(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be I_{p2}(i+w), and for switched baseband signal q1(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q2(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q1(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w), and for switched baseband signal q1(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q2(i), the inphase component may be I_{p1}(i+v) while the quadrature component may be Q_{p2}(i+w), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be I_{p2}(i+w).
 For switched baseband signal q2(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q1(i), the inphase component may be I_{p2}(i+w) while the quadrature component may be Q_{p1}(i+v).
 For switched baseband signal q2(i), the inphase component may be Q_{p2}(i+w) while the quadrature component may be I_{p1}(i+v), and for switched baseband signal q1(i), the inphase component may be Q_{p1}(i+v) while the quadrature component may be I_{p2}(i+w).
Here, weighted signal 309A(p1(i)) has an inphase component I of I_{p1}(i) and a quadrature component Q of Q_{p1}(i), while weighted signal 316B(p2(i)) has an inphase component I of I_{p2}(i) and a quadrature component Q of Q_{p2}(i). In contrast, switched baseband signal 6701A(q1(i)) has an inphase component I of I_{q1}(i) and a quadrature component Q of Q_{q1}(i), while switched baseband signal 6701B(q2(i)) has an inphase component I_{q2}(i) and a quadrature component Q of Q_{q2}(i).
In
As such, inphase component I of I_{q1}(i) and quadrature component Q of Q_{q1}(i) of switched baseband signal 6701A(q1(i)) and inphase component I_{q2}(i) and quadrature component Q of Q_{q2}(i) of baseband signal 6701B(q2(i)) are expressible as any of the above.
As such, the modulated signal corresponding to switched baseband signal 6701A(q1(i)) is transmitted from transmit antenna 312A, while the modulated signal corresponding to switched baseband signal 6701B(q2(i)) is transmitted from transmit antenna 312B, both being transmitted simultaneously on a common frequency. Thus, the modulated signals corresponding to switched baseband signal 6701A(q1(i)) and switched baseband signal 6701B(q2(i)) are transmitted from different antennas, simultaneously on a common frequency.
Phase changer 317B takes switched baseband signal 6701B and signal processing scheme information 315 as input and regularly changes the phase of switched baseband signal 6701B for output. This regular change is a change of phase performed according to a predetermined phase changing pattern having a predetermined period (cycle) (e.g., every n symbols (n being an integer, n>1) or at a predetermined interval). The phase changing pattern is described in detail in Embodiment 4.
Wireless unit 310B takes postphasechange signal 309B as input and performs processing such as quadrature modulation, band limitation, frequency conversion, amplification, and so on, then outputs transmit signal 311B. Transmit signal 311B is then output as radio waves by an antenna 312B.
Symbol 501_1 is for estimating channel fluctuations for modulated signal z2(t) (where t is time) transmitted by the transmission device. Symbol 502_1 is a data symbol transmitted by modulated signal z1(t) as symbol number u (in the time domain). Symbol 503_1 is a data symbol transmitted by modulated signal z1(t) as symbol number u+1.
Symbol 501_2 is for estimating channel fluctuations for modulated signal z2(t) (where t is time) transmitted by the transmission device. Symbol 502_2 is a data symbol transmitted by modulated signal z2(t) as symbol number u. Symbol 5032 is a data symbol transmitted by modulated signal z1(t) as symbol number u+1.
Here, the symbols of z1(t) and of z2(t) having the same time (identical timing) are transmitted from the transmit antenna using the same (shared/common) frequency.
The following describes the relationships between the modulated signals z1(t) and z2(t) transmitted by the transmission device and the received signals r1(t) and r2(t) received by the reception device.
In
Here, given vector W1=(w11,w12) from the first row of the fixed precoding matrix F, pi(t) can be expressed as formula 67, below.
[Math. 67]
p1(t)=W1s1(t) (formula 67)
Here, given vector W2=(w21,w22) from the first row of the fixed precoding matrix F, p_{2}(t) can be expressed as formula 68, below.
[Math. 68]
p2(t)=W2s2(t) (formula 68)
Accordingly, precoding matrix F may be expressed as follows.
After the baseband signals have been switched, switched baseband signal 6701A(q_{1}(i)) has an inphase component I of Iq_{1}(i) and a quadrature component Q of Qp_{1}(i), and switched baseband signal 6701B(q_{2}(i)) has an inphase component I of Iq_{2}(i) and a quadrature component Q of Qq_{2}(i). The relationships between all of these are as stated above. When the phase changer uses phase changing formula y(t), the postphasechange baseband signal 309B(q′_{2}(i)) is given by formula 70, below.
[Math. 70]
q2′(t)=y(t)q2(t) (formula 70)
Here, y(t) is a phase changing formula obeying a predetermined scheme. For example, given a period (cycle) of four and time u, the phase changing formula may be expressed as formula 71, below.
[Math. 71]
y(u)=e^{j0} (formula 71)
Similarly, the phase changing formula for time u+1 may be, for example, as given by formula 72.
That is, the phase changing formula for time u+k generalizes to formula 73.
Note that formula 71 through formula 73 are given only as an example of a regular change of phase.
The regular change of phase is not restricted to a period (cycle) of four. Improved reception capabilities (the errorcorrection capabilities, to be exact) may potentially be promoted in the reception device by increasing the period (cycle) number (this does not mean that a greater period (cycle) is better, though avoiding small numbers such as two is likely ideal).
Furthermore, although formula 71 through formula 73, above, represent a configuration in which a change of phase is carried out through rotation by consecutive predetermined phases (in the above formula, every 7/2), the change of phase need not be rotation by a constant amount but may also be random. For example, in accordance with the predetermined period (cycle) of y(t), the phase may be changed through sequential multiplication as shown in formula 74 and formula 75. The key point of the regular change of phase is that the phase of the modulated signal is regularly changed. The phase changing degree variance rate is preferably as even as possible, such as from −π radians to π radians. However, given that this concerns a distribution, random variance is also possible.
As such, the weighting unit 600 of
When a specialized precoding matrix is used in the LOS environment, the reception quality is likely to improve tremendously. However, depending on the direct wave conditions, the phase and amplitude components of the direct wave may greatly differ from the specialized precoding matrix, upon reception. The LOS environment has certain rules. Thus, data reception quality is tremendously improved through a regular change of transmit signal phase that obeys those rules. The present invention offers a signal processing scheme for improving the LOS environment.
Channel fluctuation estimator 705_1 for modulated signal z1 transmitted by the transmission device takes baseband signal 704_X as input, extracts reference symbol 501_1 for channel estimation from
Channel fluctuation estimator 705_2 for modulated signal z2 transmitted by the transmission device takes baseband signal 704_X as input, extracts reference symbol 501_2 for channel estimation from
Wireless unit 703_Y receives, as input, received signal 702_Y received by antenna 701_X, performs processing such as frequency conversion, quadrature demodulation, and the like, and outputs baseband signal 704_Y.
Channel fluctuation estimator 707_1 for modulated signal z1 transmitted by the transmission device takes baseband signal 704_Y as input, extracts reference symbol 501_1 for channel estimation from
Channel fluctuation estimator 707_2 for modulated signal z2 transmitted by the transmission device takes baseband signal 704_Y as input, extracts reference symbol 501_2 for channel estimation from
A control information decoder 709 receives baseband signal 704_X and baseband signal 704_Y as input, detects symbol 500_1 that indicates the transmission scheme from
A signal processor 711 takes the baseband signals 704_X and 704_Y, the channel estimation signals 706_1, 706_2, 708_1, and 7082, and the transmission scheme information signal 710 as input, performs detection and decoding, and then outputs received data 712_1 and 712_2.
Next, the operations of the signal processor 711 from
Accordingly, the coefficient generator 819 from
The inner MIMO detector 803 takes the signal processing scheme information signal 820 as input and performs iterative detection and decoding using the signal. The operations are described below.
The processor illustrated in
In
Subsequent operations are described separately for initial detection and for iterative decoding (iterative detection).
(Initial Detection)
The inner MIMO detector 803 takes baseband signal 801X, channel estimation signal group 802X, baseband signal 801Y, and channel estimation signal group 802Y as input. Here, the modulation scheme for modulated signal (stream) s1 and modulated signal (stream) s2 is described as 16QAM.
The inner MIMO detector 803 first computes a candidate signal point corresponding to baseband signal 801X from the channel estimation signal groups 802X and 802Y.
Similarly, the inner MIMO detector 803 calculates candidate signal points corresponding to baseband signal 801Y from channel estimation signal group 802X and channel estimation signal group 802Y, computes the Euclidean squared distance between each of the candidate signal points and the received signal points (corresponding to baseband signal 801Y), and divides the Euclidean squared distance by the noise variance 02. Accordingly, E_{Y}(b0, b1, b2, b3, b4, b5, b6, b7) is calculated. That is, E_{Y }is the Euclidian squared distance between a candidate signal point corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and a received signal point, divided by the noise variance.
Next, E_{X}(b0, b1, b2, b3, b4, b5, b6, b7)+E_{Y}(b0, b1, b2, b3, b4, b5, b6, b7)=E(b0, b1, b2, b3, b4, b5, b6, b7) is computed.
The inner MIMO detector 803 outputs E(b0, b1, b2, b3, b4, b5, b6, b7) as the signal 804.
The loglikelihood calculator 805A takes the signal 804 as input, calculates the loglikelihood of bits b0, b1, b2, and b3, and outputs the loglikelihood signal 806A. Note that this loglikelihood calculation produces the loglikelihood of a bit being 1 and the loglikelihood of a bit being 0. The calculation is as shown in formula 28, formula 29, and formula 30, and the details thereof are given by NonPatent Literature 2 and 3.
Similarly, loglikelihood calculator 805B takes the signal 804 as input, calculates the loglikelihood of bits b4, b5, b6, and b7, and outputs loglikelihood signal 806A.
A deinterleaver (807A) takes loglikelihood signal 806A as input, performs deinterleaving corresponding to that of the interleaver (the interleaver (304A) from
Similarly, a deinterleaver (807B) takes loglikelihood signal 806B as input, performs deinterleaving corresponding to that of the interleaver (the interleaver (304B) from
Loglikelihood ratio calculator 809A takes deinterleaved loglikelihood signal 808A as input, calculates the loglikelihood ratio of the bits encoded by encoder 302A from
Similarly, loglikelihood ratio calculator 809B takes deinterleaved loglikelihood signal 808B as input, calculates the loglikelihood ratio of the bits encoded by encoder 302B from
Softin/softout decoder 811A takes loglikelihood ratio signal 810A as input, performs decoding, and outputs a decoded loglikelihood ratio 812A.
Similarly, softin/softout decoder 811B takes loglikelihood ratio signal 810B as input, performs decoding, and outputs decoded loglikelihood ratio 812B.
(Iterative Decoding (Iterative Detection), k Iterations)
The interleaver (813A) takes the k−1th decoded loglikelihood ratio 812A decoded by the softin/softout decoder as input, performs interleaving, and outputs an interleaved loglikelihood ratio 814A. Here, the interleaving pattern used by the interleaver (813A) is identical to that of the interleaver (304A) from
Another interleaver (813B) takes the k−1th decoded loglikelihood ratio 812B decoded by the softin/softout decoder as input, performs interleaving, and outputs interleaved loglikelihood ratio 814B. Here, the interleaving pattern used by the interleaver (813B) is identical to that of the other interleaver (304B) from
The inner MIMO detector 803 takes baseband signal 816X, transformed channel estimation signal group 817X, baseband signal 816Y, transformed channel estimation signal group 817Y, interleaved loglikelihood ratio 814A, and interleaved loglikelihood ratio 814B as input. Here, baseband signal 816X, transformed channel estimation signal group 817X, baseband signal 816Y, and transformed channel estimation signal group 817Y are used instead of baseband signal 801X, channel estimation signal group 802X, baseband signal 801Y, and channel estimation signal group 802Y because the latter cause delays due to the iterative decoding.
The iterative decoding operations of the inner MIMO detector 803 differ from the initial detection operations thereof in that the interleaved loglikelihood ratios 814A and 814B are used in signal processing for the former. The inner MIMO detector 803 first calculates E(b0, b1, b2, b3, b4, b5, b6, b7) in the same manner as for initial detection. In addition, the coefficients corresponding to formula 11 and formula 32 are computed from the interleaved loglikelihood ratios 814A and 914B. The value of E(b0, b1, b2, b3, b4, b5, b6, b7) is corrected using the coefficients so calculated to obtain E′(b0, b1, b2, b3, b4, b5, b6, b7), which is output as the signal 804.
Loglikelihood calculator 805A takes the signal 804 as input, calculates the loglikelihood of bits b0, b1, b2, and b3, and outputs a loglikelihood signal 806A. Note that this loglikelihood calculation produces the loglikelihood of a bit being 1 and the loglikelihood of a bit being 0. The calculation is as shown in formula 31 through formula 35, and the details are given by NonPatent Literature 2 and 3.
Similarly, loglikelihood calculator 805B takes the signal 804 as input, calculates the loglikelihood of bits b4, b5, b6, and b7, and outputs loglikelihood signal 806B. Operations performed by the deinterleaver onwards are similar to those performed for initial detection.
While
As shown in NonPatent Literature 5 and the like, QR decomposition may also be used to perform initial detection and iterative detection. Also, as indicated by NonPatent Literature 11, MMSE and ZF linear operations may be performed when performing initial detection.
As described above, when a transmission device according to the present embodiment using a MIMO system transmits a plurality of modulated signals from a plurality of antennas, changing the phase over time while multiplying by the precoding matrix so as to regularly change the phase results in improvements to data reception quality for a reception device in a LOS environment, where direct waves are dominant, compared to a conventional spatial multiplexing MIMO system.
In the present embodiment, and particularly in the configuration of the reception device, the number of antennas is limited and explanations are given accordingly. However, the Embodiment may also be applied to a greater number of antennas. In other words, the number of antennas in the reception device does not affect the operations or advantageous effects of the present embodiment.
Further, in the present embodiments, the encoding is not particularly limited to LDPC codes. Similarly, the decoding scheme is not limited to implementation by a softin/softout decoder using sumproduct decoding. The decoding scheme used by the softin/softout decoder may also be, for example, the BCJR algorithm, SOVA, and the MaxLogMap algorithm. Details are provided in NonPatent Literature 6.
In addition, although the present embodiment is described using a singlecarrier scheme, no limitation is intended in this regard. The present embodiment is also applicable to multicarrier transmission. Accordingly, the present embodiment may also be realized using, for example, spreadspectrum communications, OFDM, SCFDMA, SCOFDM, wavelet OFDM as described in NonPatent Literature 7, and so on. Furthermore, in the present embodiment, symbols other than data symbols, such as pilot symbols (preamble, unique word, and so on) or symbols transmitting control information, may be arranged within the frame in any manner.
The following describes an example in which OFDM is used as a multicarrier scheme.
An OFDMrelated processor 1201A takes weighted signal 309A as input, performs OFDMrelated processing thereon, and outputs transmit signal 1202A. Similarly, OFDMrelated processor 1201B takes postphasechange signal 309B as input, performs OFDMrelated processing thereon, and outputs transmit signal 1202B.
Serialtoparallel converter 1302A performs serialtoparallel conversion on switched baseband signal 1301A (corresponding to switched baseband signal 6701A from
Reorderer 1304A takes parallel signal 1303A as input, performs reordering thereof, and outputs reordered signal 1305A. Reordering is described in detail later.
IFFT unit 1306A takes reordered signal 1305A as input, applies an IFFT thereto, and outputs postIFFT signal 1307A.
Wireless unit 1308A takes postIFFT signal 1307A as input, performs processing such as frequency conversion and amplification, thereon, and outputs modulated signal 1309A. Modulated signal 1309A is then output as radio waves by antenna 1310A.
Serialtoparallel converter 1302B performs serialtoparallel conversion on postphasechange signal 1301B (corresponding to postphasechange signal 309B from
Reorderer 1304B takes parallel signal 1303B as input, performs reordering thereof, and outputs reordered signal 1305B. Reordering is described in detail later.
IFFT unit 1306B takes reordered signal 1305B as input, applies an IFFT thereto, and outputs postIFFT signal 1307B.
Wireless unit 1308B takes postIFFT signal 1307B as input, performs processing such as frequency conversion and amplification thereon, and outputs modulated signal 1309B. Modulated signal 1309B is then output as radio waves by antenna 1310A.
The transmission device from
As shown in
Similarly, with respect to the symbols of weighted signal 1301B input to serialtoparallel converter 1302B, the assigned ordering is #0, #1, #2, #3, and so on. Here, given that the example deals with a period (cycle) of four, a different change in phase is applied to each of #0, #1, #2, and #3, which are equivalent to one period (cycle). Similarly, a different change in phase is applied to each of #4n, #4n+1, #4n+2, and #4n+3 (n being a nonzero positive integer), which are also equivalent to one period (cycle)
As shown in
The symbol group 1402 shown in
In the present embodiment, modulated signal z1 shown in
As such, when using a multicarrier transmission scheme such as OFDM, and unlike single carrier transmission, symbols can be arranged in the frequency domain. Of course, the symbol arrangement scheme is not limited to those illustrated by
While
In
Here, symbol #0 is obtained using the change of phase at time u, symbol #1 is obtained using the change of phase at time u+1, symbol #2 is obtained using the change of phase at time u+2, and symbol #3 is obtained using the change of phase at time u+3.
Similarly, for frequencydomain symbol group 2220, symbol #4 is obtained using the change of phase at time u, symbol #5 is obtained using the change of phase at time u+1, symbol #6 is obtained using the change of phase at time u+2, and symbol #7 is obtained using the change of phase at time u+3.
The abovedescribed change of phase is applied to the symbol at time $1. However, in order to apply periodic shifting with respect to the time domain, the following change of phases are applied to symbol groups 2201, 2202, 2203, and 2204.
For timedomain symbol group 2201, symbol #0 is obtained using the change of phase at time u, symbol #9 is obtained using the change of phase at time u+1, symbol #18 is obtained using the change of phase at time u+2, and symbol #27 is obtained using the change of phase at time u+3.
For timedomain symbol group 2202, symbol #28 is obtained using the change of phase at time u, symbol #1 is obtained using the change of phase at time u+1, symbol #10 is obtained using the change of phase at time u+2, and symbol #19 is obtained using the change of phase at time u+3.
For timedomain symbol group 2203, symbol #20 is obtained using the change of phase at time u, symbol #29 is obtained using the change of phase at time u+1, symbol #2 is obtained using the change of phase at time u+2, and symbol #11 is obtained using the change of phase at time u+3.
For timedomain symbol group 2204, symbol #12 is obtained using the change of phase at time u, symbol #21 is obtained using the change of phase at time u+1, symbol #30 is obtained using the change of phase at time u+2, and symbol #3 is obtained using the change of phase at time u+3.
The characteristic feature of
Although
Although the present embodiment describes a variation of Embodiment 1 in which a baseband signal switcher is inserted before the change of phase, the present embodiment may also be realized as a combination with Embodiment 2, such that the baseband signal switcher is inserted before the change of phase in
The following describes a scheme for allowing the reception device to obtain good received signal quality for data, regardless of the reception device arrangement, by considering the location of the reception device with respect to the transmission device.
Consider symbol 3100 at carrier 2 and time $2 of
Within carrier 2, there is a very strong correlation between the channel conditions for symbol 610A at carrier 2, time $2 and the channel conditions for the time domain nearestneighbour symbols to time $2, i.e., symbol 3013 at time $1 and symbol 3101 at time $3 within carrier 2.
Similarly, for time $2, there is a very strong correlation between the channel conditions for symbol 3100 at carrier 2, time $2 and the channel conditions for the frequencydomain nearestneighbour symbols to carrier 2, i.e., symbol 3104 at carrier 1, time $2 and symbol 3104 at time $2, carrier 3.
As described above, there is a very strong correlation between the channel conditions for symbol 3100 and the channel conditions for each symbol 3101, 3102, 3103, and 3104.
The present description considers N different phases (N being an integer, N>2) for multiplication in a transmission scheme where the phase is regularly changed. The symbols illustrated in
The present embodiment takes advantage of the high correlation in channel conditions existing between neighbouring symbols in the frequency domain and/or neighbouring symbols in the time domain in a symbol arrangement enabling high data reception quality to be obtained by the reception device receiving the postphasechange symbols.
In order to achieve this high data reception quality, conditions #D11 and #D12 should preferably be met.
As shown in
As shown in
Ideally, a data symbol should satisfy Condition #D11. Similarly, the data symbols should satisfy Condition #D12.
The reasons supporting Conditions #D11 and #D12 are as follows.
A very strong correlation exists between the channel conditions of given symbol of a transmit signal (hereinafter, symbol A) and the channel conditions of the symbols neighbouring symbol A in the time domain, as described above.
Accordingly, when three neighbouring symbols in the time domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to phase relations despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding.
Similarly, a very strong correlation exists between the channel conditions of given symbol of a transmit signal (symbol A) and the channel conditions of the symbols neighbouring symbol A in the frequency domain, as described above.
Accordingly, when three neighbouring symbols in the frequency domain each have different phases, then despite reception quality degradation in the LOS environment (poor signal quality caused by degradation in conditions due to direct wave phase relationships despite high signal quality in terms of SNR) for symbol A, the two remaining symbols neighbouring symbol A are highly likely to provide good reception quality. As a result, good received signal quality is achievable after error correction and decoding.
Combining Conditions #D11 and #D12, ever greater data reception quality is likely achievable for the reception device. Accordingly, the following Condition #D13 can be derived.
As shown in
Here, the different changes in phase are as follows. Phase changes are defined from 0 radians to 271 radians. For example, for time X, carrier Y, a phase change of e^{jθX,Y }is applied to precoded baseband signal q_{2 }from
Ideally, a data symbol should satisfy Condition #D11.
As evident from
In other words, in
Similarly, in
Similarly, in
The following discusses the abovedescribed example for a case where the change of phase is performed on two switched baseband signals q1 and q2 (see
Several phase changing schemes are applicable to performing a change of phase on two switched baseband signals q1 and q2. The details thereof are explained below.
Scheme 1 involves a change of phase of switched baseband signal q2 as described above, to achieve the change of phase illustrated by
The symbols illustrated in
As shown in
As described above, the change in phase performed on switched baseband signal q2 has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the degree of phase change applied to switched baseband signal q1 and to switched baseband signal q2 into consideration. Accordingly, data reception quality may be improved for the reception device.
Scheme 2 involves a change in phase of switched baseband signal q2 as described above, to achieve the change in phase illustrated by
The symbols illustrated in
As described above, the change in phase performed on switched baseband signal q_{2 }has a period (cycle) of ten, but the period (cycle) can be effectively made greater than ten by taking the changes in phase applied to switched baseband signal ql and to switched baseband signal q2 into consideration. Accordingly, data reception quality may be improved for the reception device. An effective way of applying scheme 2 is to perform a change in phase on switched baseband signal ql with a period (cycle) of N and perform a change in phase on precoded baseband signal q2 with a period (cycle) of M such that N and M are coprime. As such, by taking both switched baseband signals q1 and q2 into consideration, a period (cycle) of N×M is easily achievable, effectively making the period (cycle) greater when N and M are coprime.
While the above discusses an example of the abovedescribed phase changing scheme, the present invention is not limited in this manner. The change in phase may be performed with respect to the frequency domain, the time domain, or on timefrequency blocks. Similar improvement to the data reception quality can be obtained for the reception device in all cases.
The same also applies to frames having a configuration other than that described above, where pilot symbols (SP symbols) and symbols transmitting control information are inserted among the data symbols. The details of the change in phase in such circumstances are as follows.
The important point of
The important point of
The important point of
The important point of
In
In
Although not indicated in the frame configurations from
The wireless units 310A and 310B of
A selector 5301 takes the plurality of baseband signals as input and selects a baseband signal having a symbol indicated by the frame configuration signal 313 for output.
Similarly, as shown in
The above explanations are given using pilot symbols, control symbols, and data symbols as examples. However, the present invention is not limited in this manner. When symbols are transmitted using schemes other than precoding, such as singleantenna transmission or transmission using spacetime block codes, the absence of change in phase is important. Conversely, performing the change of phase on symbols that have been precoded is the key point of the present invention.
Accordingly, a characteristic feature of the present invention is that the change in phase is not performed on all symbols within the frame configuration in the timefrequency domain, but only performed on baseband signals that have been precoded and have undergone switching.
The following describes a scheme for regularly changing the phase when encoding is performed using block codes as described in NonPatent Literature 12 through 15, such as QC LDPC Codes (not only QCLDPC but also LDPC codes may be used), concatenated LDPC and BCH codes, Turbo codes or DuoBinary Turbo Codes using tailbiting, and so on. The following example considers a case where two streams s1 and s2 are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is necessary, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information.
As shown in
Then, given that the abovedescribed transmission device transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s1 and the other 1500 symbols are assigned to s2. As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s1 and s2.
By the same reasoning, when the modulation scheme is 16QAM, 750 slots are needed to transmit all of the bits making up one coded block, and when the modulation scheme is 64QAM, 500 slots are needed to transmit all of the bits making up one coded block.
The following describes the relationship between the abovedefined slots and the phase of multiplication, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase. That is, the phase changer of the abovedescribed transmission device uses five phase changing values (or phase changing sets) to achieve the period (cycle) of five. (As in
For the abovedescribed 1500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is QPSK, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality.
Furthermore, for the abovedescribed 750 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 16QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 150 slots, PHASE[2] is used on 150 slots, PHASE[3] is used on 150 slots, and PHASE[4] is used on 150 slots.
Further still, for the abovedescribed 500 slots needed to transmit the 6000 bits making up a single coded block when the modulation scheme is 64QAM, PHASE[0] is used on 150 slots, PHASE[1] is used on 100 slots, PHASE[2] is used on 100 slots, PHASE[3] is used on 100 slots, and PHASE[4] is used on 100 slots.
As described above, a scheme for a regular change of phase requires the preparation of N phase changing values (or phase changing sets) (where the N different phases are expressed as PHASE[0], PHASE[1], PHASE[2], PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bits making up a single coded block, PHASE[0] is used on K_{0 }slots, PHASE[1] is used on K_{1 }slots, PHASE[i] is used on K_{i }slots (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1] is used on K_{N−1 }slots, such that Condition #D14 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K N−1. That is, K_{a}=K_{b }(for ∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Then, when a communication system that supports multiple modulation schemes selects one such supported scheme for use, Condition #D14 is preferably satisfied for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #D14 may not be satisfied for some modulation schemes. In such a case, the following condition applies instead of Condition #D14.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
As shown in
The transmission device from
By the same reasoning, when the modulation scheme is 16QAM, 1500 slots are needed to transmit all of the bits making up the two coded blocks, and when the modulation scheme is 64QAM, 1000 slots are needed to transmit all of the bits making up the two coded blocks.
The following describes the relationship between the abovedefined slots and the phase of multiplication, as pertains to schemes for a regular change of phase.
Here, five different phase changing values (or phase changing sets) are assumed as having been prepared for use in the scheme for a regular change of phase. That is, the phase changer of the transmission device from
For the abovedescribed 3000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is QPSK, PHASE[0] is used on 600 slots, PHASE[1] is used on 600 slots, PHASE[2] is used on 600 slots, PHASE[3] is used on 600 slots, and PHASE[4] is used on 600 slots. This is due to the fact that any bias in phase usage causes great influence to be exerted by the more frequently used phase, and that the reception device is dependent on such influence for data reception quality.
Further, in order to transmit the first coded block, PHASE[0] is used on slots 600 times, PHASE[1] is used on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] is used on slots 600 times. Furthermore, in order to transmit the second coded block, PHASE[0] is used on slots 600 times, PHASE[1] is used on slots 600 times, PHASE[2] is used on slots 600 times, PHASE[3] is used on slots 600 times, and PHASE[4] is used on slots 600 times.
Similarly, for the abovedescribed 1500 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 16QAM, PHASE[0] is used on 300 slots, PHASE[1] is used on 300 slots, PHASE[2] is used on 300 slots, PHASE[3] is used on 300 slots, and PHASE[4] is used on 300 slots.
Further, in order to transmit the first coded block, PHASE[0] is used on slots 300 times, PHASE[1] is used on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] is used on slots 300 times. Furthermore, in order to transmit the second coded block, PHASE[0] is used on slots 300 times, PHASE[1] is used on slots 300 times, PHASE[2] is used on slots 300 times, PHASE[3] is used on slots 300 times, and PHASE[4] is used on slots 300 times.
Similarly, for the abovedescribed 1000 slots needed to transmit the 6000×2 bits making up the two coded blocks when the modulation scheme is 64QAM, PHASE[0] is used on 200 slots, PHASE[1] is used on 200 slots, PHASE[2] is used on 200 slots, PHASE[3] is used on 200 slots, and PHASE[4] is used on 200 slots.
Further, in order to transmit the first coded block, PHASE[0] is used on slots 200 times, PHASE[1] is used on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] is used on slots 200 times. Furthermore, in order to transmit the second coded block, PHASE[0] is used on slots 200 times, PHASE[1] is used on slots 200 times, PHASE[2] is used on slots 200 times, PHASE[3] is used on slots 200 times, and PHASE[4] is used on slots 200 times.
As described above, a scheme for a regular change of phase requires the preparation of N phase changing values (or phase changing sets) (where the N different phases are expressed as PHASE[0], PHASE[1], PHASE[2], PHASE[N−2], PHASE[N−1]). As such, in order to transmit all of the bits making up a single coded block, PHASE[0] is used on K_{0 }slots, PHASE[1] is used on K_{1 }slots, PHASE[i] is used on K_{i }slots (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1] is used on K_{N−1 }slots, such that Condition #D16 is met.
K_{0}=K_{1 }. . . =K_{i}= . . . K_{N−1}. That is, K_{a}=K_{b }(for ∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a b).
Further, in order to transmit all of the bits making up the first coded block, PHASE[0] is used K_{0,1 }times, PHASE[1] is used K_{1,1 }times, PHASE[i] is used K_{i,1 }times (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1] is used K_{N−1,1 }times, such that Condition #D17 is met.
K_{0,1}=K_{1,1}= . . . K_{i,1}= . . . K_{N−1,1}. That is, K_{a},1=K_{b,1 }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Furthermore, in order to transmit all of the bits making up the second coded block, PHASE[0] is used K_{0,2 }times, PHASE[1] is used K_{1,2 }times, PHASE[i] is used K_{i,2 }times (where i=0, 1, 2, . . . , N−1 (i denotes an integer that satisfies 0≤i≤N−1)), and PHASE[N−1] is used K_{N−1,2 }times, such that Condition #D18 is met.
K_{0,2}=K_{1,2}= . . . K_{i,2}= . . . K_{N−1,2}. That is, K_{a},2=K_{b,2 }(∀a and ∀b where a, b, =0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b).
Then, when a communication system that supports multiple modulation schemes selects one such supported scheme for use, Condition #D16 Condition #D17, and Condition #D18 are preferably satisfied for the supported modulation scheme.
However, when multiple modulation schemes are supported, each such modulation scheme typically uses symbols transmitting a different number of bits per symbols (though some may happen to use the same number), Condition #D16 Condition #D17, and Condition #D18 may not be satisfied for some modulation schemes. In such a case, the following conditions apply instead of Condition #D16 Condition #D17, and Condition #D18.
The difference between K_{a }and K_{b }satisfies 0 or 1. That is, K_{a}K_{b }satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
The difference between K_{a,1 }and K_{b,1 }satisfies 0 or 1. That is, K_{a,1}−K_{b,1}l satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
The difference between K_{a,2 }and K_{b,2 }satisfies 0 or 1. That is, K_{a,2}−K_{b,2} satisfies 0 or 1 (∀a, ∀b, where a, b=0, 1, 2, . . . , N−1 (a denotes an integer that satisfies 0≤a≤N−1, b denotes an integer that satisfies 0≤b≤N−1), a≠b)
As described above, bias among the phases being used to transmit the coded blocks is removed by creating a relationship between the coded block and the phase of multiplication. As such, data reception quality may be improved for the reception device.
As described above, N phase changing values (or phase changing sets) are needed in order to perform a change of phase having a period (cycle) of N with the scheme for the regular change of phase. As such, N phase changing values (or phase changing sets) PHASE[0], PHASE[1], PHASE[2], . . . , PHASE[N−2], and PHASE[N−1] are prepared. However, schemes exist for ordering the phases in the stated order with respect to the frequency domain. No limitation is intended in this regard. The N phase changing values (or phase changing sets) PHASE[0], PHASE[1], PHASE[2], . . . , PHASE[N−2], and PHASE[N−1] may also change the phases of blocks in the time domain or in the timefrequency domain to obtain a symbol arrangement. Although the above examples discuss a phase changing scheme with a period (cycle) of N, the same effects are obtainable using N phase changing values (or phase changing sets) at random. That is, the N phase changing values (or phase changing sets) need not always have regular periodicity. As long as the abovedescribed conditions are satisfied, great quality data reception improvements are realizable for the reception device.
Furthermore, given the existence of modes for spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase, the transmission device (broadcaster, base station) may select any one of these transmission schemes.
As described in NonPatent Literature 3, spatial multiplexing MIMO schemes involve transmitting signals s1 and s2, which are mapped using a selected modulation scheme, on each of two different antennas. MIMO schemes using a fixed precoding matrix involve performing precoding only (with no change in phase). Further, spacetime block coding schemes are described in NonPatent Literature 9, 16, and 17. Singlestream transmission schemes involve transmitting signal s1, mapped with a selected modulation scheme, from an antenna after performing predetermined processing.
Schemes using multicarrier transmission such as OFDM involve a first carrier group made up of a plurality of carriers and a second carrier group made up of a plurality of carriers different from the first carrier group, and so on, such that multicarrier transmission is realized with a plurality of carrier groups. For each carrier group, any of spatial multiplexing MIMO schemes, MIMO schemes using a fixed precoding matrix, spacetime block coding schemes, singlestream transmission, and schemes using a regular change of phase may be used. In particular, schemes using a regular change of phase on a selected (sub)carrier group are preferably used to realize the above.
Although the present description describes the present embodiment as a transmission device applying precoding, baseband switching, and change in phase, all of these may be variously combined. In particular, the phase changer discussed for the present embodiment may be freely combined with the change in phase discussed in all other Embodiments.
The present embodiment describes a phase change initialization scheme for the regular change of phase described throughout the present description. This initialization scheme is applicable to the transmission device from
The following is also applicable to a scheme for regularly changing the phase when encoding is performed using block codes as described in NonPatent Literature 12 through 15, such as QC LDPC Codes (not only QCLDPC but also LDPC codes may be used), concatenated LDPC and BCH codes, Turbo codes or DuoBinary Turbo Codes using tailbiting, and so on.
The following example considers a case where two streams s1 and s2 are transmitted. When encoding has been performed using block codes and control information and the like is not necessary, the number of bits making up each coded block matches the number of bits making up each block code (control information and so on described below may yet be included). When encoding has been performed using block codes or the like and control information or the like (e.g., CRC transmission parameters) is required, then the number of bits making up each coded block is the sum of the number of bits making up the block codes and the number of bits making up the information.
As shown in
Then, given that the abovedescribed transmission device transmits two streams simultaneously, 1500 of the aforementioned 3000 symbols needed when the modulation scheme is QPSK are assigned to s1 and the other 1500 symbols are assigned to s2. As such, 1500 slots for transmitting the 1500 symbols (hereinafter, slots) are required for each of s1 and s2.
By the same reasoning, when the modulation scheme is 16QAM, 750 slots are needed to transmit all of the bits making up each coded block, and when the modulation scheme is 64QAM, 500 slots are needed to transmit all of the bits making up each coded block.
The following describes a transmission device transmitting modulated signals having a frame configuration illustrated by
As shown in
Further, the transmission device transmits a preamble (control symbol) during interval D. The preamble is a symbol transmitting control information for another party. In particular, this preamble includes information on the modulation scheme used to transmit a third or fourth coded block and so on. The transmission device transmits the third coded block during interval E. The transmission device then transmits the fourth coded block during interval D.
Also, as shown in
Further, the transmission device transmits a preamble (control symbol) during interval D. The preamble is a symbol transmitting control information for another party. In particular, this preamble includes information on the modulation scheme used to transmit a third or fourth coded block and so on. The transmission device transmits the third coded block during interval E. The transmission device then transmits the fourth coded block during interval D.
Similarly,
As explained throughout this description, modulated signal z1, i.e., the modulated signal transmitted by antenna 312A, does not undergo a change in phase, while modulated signal z2, i.e., the modulated signal transmitted by antenna 312B, does undergo a change in phase. The following phase changing scheme is used for
Before the change in phase can occur, seven different phase changing values is prepared. The seven phase changing values are labeled #0, #1, #2, #3, #4, #5, #6, and #7. The change in phase is regular and periodic. In other words, the phase changing values are applied regularly and periodically, such that the order is #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6 and so on.
As shown in
The change in phase is then applied to each slot for the second coded block. The present description assumes multicast transmission and broadcasting applications. As such, a receiving terminal may have no need for the first coded block and extract only the second coded block. In such circumstances, given that the final slot used for the first coded block uses phase changing value #0, the initial phase changing value used for the second coded block is #1. As such, the following schemes are conceivable:
(a): The aforementioned terminal monitors the transmission of the first coded block, i.e., monitors the pattern of the phase changing values through the final slot used to transmit the first coded block, and then estimates the phase changing value used for the initial slot of the second coded block;
(b): (a) does not occur, and the transmission device transmits information on the phase changing values in use at the initial slot of the second coded block. Scheme (a) leads to greater energy consumption by the terminal due to the need to monitor the transmission of the first coded block. However, scheme (b) leads to reduced data transmission efficiency.
Accordingly, there is a need to improve the phase changing value allocation described above. Consider a scheme in which the phase changing value used to transmit the initial slot of each coded block is fixed. Thus, as indicated in
Similarly, as indicated in
As such, the problems accompanying both schemes (a) and (b) described above can be constrained while retaining the effects thereof.
In the present embodiment, the scheme used to initialize the phase changing value for each coded block, i.e., the phase changing value used for the initial slot of each coded block, is fixed so as to be #0. However, other schemes may also be used for singleframe units. For example, the phase changing value used for the initial slot of a symbol transmitting information after the preamble or control symbol has been transmitted may be fixed at #0.
The abovedescribed Embodiments discuss a weighting unit using a precoding matrix expressed in complex numbers for precoding. However, the precoding matrix may also be expressed in real numbers.
That is, suppose that two baseband signals s_{1}(i) and s2(i) (where i is time or frequency) have been mapped (using a modulation scheme), and precoded to obtained precoded baseband signals z1(i) and z2(i). As such, mapped baseband signal s_{1}(i) has an inphase component of I_{s1}(i) and a quadrature component of Q_{s1}(i), and mapped baseband signal s2(i) has an inphase component of Is_{2}(i) and a quadrature component of Q_{s2}(i), while precoded baseband signal z1(i) has an inphase component of Iz1(i) and a quadrature component of Q_{z1}(i), and precoded baseband signal z2(i) has an inphase component of I_{z2}(i) and a quadrature component of Q_{z2}(i), which gives the following precoding matrix H_{r }when all values are real numbers.
Precoding matrix H_{r }may also be expressed as follows, where all values are real numbers.
where a_{11}, a_{12}, a_{13}, a_{14}, a_{21}, a_{22}, a_{23}, a_{24}, a_{31}, a_{32}, a_{33}, a_{34}, a_{41}, a_{42}, a_{43}, and a_{44 }are real numbers. However, none of the following may hold: {a_{11}=0, a_{12}=0, a_{13}=0, and a_{14}=0}, {a_{21}=0, a_{22}=0, a_{23}=0, and a_{24}=0}, {a_{31}=0, a_{32}=0, a_{33}=0, and a_{34}=0}, and {a_{41}=0, a_{42}=0, a_{43}=0, and a_{44}=0}. Also, none of the following may hold: {a_{11}=0, a_{21}=0, a_{31}=0, and a_{41}=0}, {a_{12}=0, a_{22}=0, a_{32}=0, and a_{42}=0}, {a_{13}=0, a_{23}=0, a_{33}=0, and a_{43}=0}, and {a_{14}=0, a_{24}=0, a_{34}=0, and a_{44}=0}.
The present embodiment describes a scheme of initializing phase change in a case where (i) the transmission device in
The following describes the scheme for regularly changing the phase when using a QuasiCyclic LowDensity ParityCheck (QCLDPC) code (or an LDPC code other than a QCLDPC code), a concatenated code consisting of an LDPC code and a BoseChaudhuriHocquenghem (BCH) code, and a block code such as a turbo code or a duobinary turbo code using tailbiting. These codes are described in NonPatent Literatures 12 through 15.
The following describes a case of transmitting two streams s1 and s2 as an example. Note that, when the control information and the like are not required to perform encoding using the block code, the number of bits constituting the coding (encoded) block is the same as the number of bits constituting the block code (however, the control information and the like described below may be included). When the control information and the like (e.g. CRC (cyclic redundancy check), a transmission parameter) are required to perform encoding using the block code, the number of bits constituting the coding (encoded) block can be a sum of the number of bits constituting the block code and the number of bits of the control information and the like.
As shown in
Since two streams are to be simultaneously transmitted in the transmission device above, when the modulation scheme is QPSK, 1500 symbols are allocated to s1 and remaining 1500 symbols are allocated to s2 out of the abovementioned 3000 symbols. Therefore, 1500 slots (referred to as slots) are necessary to transmit 1500 symbols by s1 and transmit 1500 symbols by s2.
Making the same considerations, 750 slots are necessary to transmit all the bits constituting one coding (encoded) block when the modulation scheme is 16QAM, and 500 slots are necessary to transmit all the bits constituting one block when the modulation scheme is 64QAM.
Next, a case where the transmission device transmits modulated signals each having a frame structure shown in
As shown in
The transmission device transmits the preamble (control symbol) in an interval D. The preamble is a symbol for transmitting control information to the communication partner and is assumed to include information on the modulation scheme for transmitting the third coding (encoded) block, the fourth coding (encoded) block and so on. The transmission device is to transmit the third coding (encoded) block in an interval E. The transmission device is to transmit the fourth coding (encoded) block in an interval F.
As shown in
The transmission device transmits the preamble (control symbol) in the interval D. The preamble is a symbol for transmitting control information to the communication partner and is assumed to include information on the modulation scheme for transmitting the third coding (encoded) block, the fourth coding (encoded) block and so on. The transmission device is to transmit the third coding (encoded) block in the interval E. The transmission device is to transmit the fourth coding (encoded) block in the interval F.
Similarly,
As described in this description, a case where phase change is not performed for the modulated signal z1, i.e. the modulated signal transmitted by the antenna 312A, and is performed for the modulated signal z2, i.e. the modulated signal transmitted by the antenna 312B, is considered. In this case,
First, assume that seven different phase changing values are prepared to perform phase change, and are referred to as #0, #1, #2, #3, #4, #5 and #6. The phase changing values are to be regularly and cyclically used. That is to say, the phase changing values are to be regularly and cyclically changed in the order such as #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, . . . .
First, as shown in
Next, the phase changing values are to be applied to each slot in the second coding (encoded) block. Since this description is on the assumption that the phase changing values are applied to the multicast communication and broadcast, one possibility is that a reception terminal does not need the first coding (encoded) block and extracts only the second coding (encoded) block. In such a case, even when phase changing value #0 is used to transmit the last slot in the first coding (encoded) block, the phase changing value #1 is used first to transmit the second coding (encoded) block. In this case, the following two schemes are considered:
(a) The abovementioned terminal monitors how the first coding (encoded) block is transmitted, i.e. the terminal monitors a pattern of the phase changing value used to transmit the last slot in the first coding (encoded) block, and estimates the phase changing value to be used to transmit the first slot in the second coding (encoded) block; and
(b) The transmission device transmits information on the phase changing value used to transmit the first slot in the second coding (encoded) block without performing (a).
In the case of (a), since the terminal has to monitor transmission of the first coding (encoded) block, power consumption increases. In the case of (b), transmission efficiency of data is reduced.
Therefore, there is room for improvement in allocation of precoding matrices as described above. In order to address the abovementioned problems, a scheme of fixing the phase changing value used to transmit the first slot in each coding (encoded) block is proposed. Therefore, as shown in
Similarly, as shown in
With the abovementioned scheme, an effect of suppressing the problems occurring in (a) and (b) is obtained.
Note that, in the present embodiment, the scheme of initializing the phase changing values in each coding (encoded) block, i.e. the scheme in which the phase changing value used to transmit the first slot in each coding (encoded) block is fixed to #0, is described. As a different scheme, however, the phase changing values may be initialized in units of frames. For example, in the symbol for transmitting the preamble and information after transmission of the control symbol, the phase changing value used in the first slot may be fixed to #0.
For example, in
The following describes a case where the abovementioned scheme is applied to a broadcasting system that uses the DVBT2 standard. First, the frame structure for a broadcast system according to the DVBT2 standard is described.
The P1 Signalling data (7401) is a symbol for use by a reception device for signal detection and frequency synchronization (including frequency offset estimation). Also, the P1 Signalling data (7401) transmits information including information indicating the FFT (Fast Fourier Transform) size, and information indicating which of SISO (SingleInput SingleOutput) and MISO (MultipleInput SingleOutput) is employed to transmit a modulated signal. (The SISO scheme is for transmitting one modulated signal, whereas the MISO scheme is for transmitting a plurality of modulated signals using spacetime block codes shown in NonPatent Literatures 9, 16 and 17.)
The L1 PreSignalling data (7402) transmits information including: information about the guard interval used in transmitted frames; information about the signal processing method for reducing PAPR (Peak to Average Power Ratio); information about the modulation scheme, error correction scheme (FEC: Forward Error Correction), and coding rate of the error correction scheme all used in transmitting L1 PostSignalling data; information about the size of L1 PostSignalling data and the information size; information about the pilot pattern; information about the cell (frequency region) unique number; and information indicating which of the normal mode and extended mode (the respective modes differs in the number of subcarriers used in data transmission) is used.
The L1 PostSignalling data (7403) transmits information including: information about the number of PLPs; information about the frequency region used; information about the unique number of each PLP; information about the modulation scheme, error correction scheme, coding rate of the error correction scheme all used in transmitting the PLPs; and information about the number of blocks transmitted in each PLP.
The Common PLP (7404) and PLPs #1 to #N (7405_1 to 7405_N) are fields used for transmitting data.
In the frame structure shown in
A PLP signal generator 7602 receives PLP transmission data (transmission data for a plurality of PLPs) 7601 and a control signal 7609 as input, performs mapping of each PLP according to the error correction scheme and modulation scheme indicated for the PLP by the information included in the control signal 7609, and outputs a (quadrature) baseband signal 7603 carrying a plurality of PLPs.
A P2 symbol signal generator 7605 receives P2 symbol transmission data 7604 and the control signal 7609 as input, performs mapping according to the error correction scheme and modulation scheme indicated for each P2 symbol by the information included in the control signal 7609, and outputs a (quadrature) baseband signal 7606 carrying the P2 symbols.
A control signal generator 7608 receives P1 symbol transmission data 7607 and P2 symbol transmission data 7604 as input, and then outputs, as the control signal 7609, information about the transmission scheme (the error correction scheme, coding rate of the error correction, modulation scheme, block length, frame structure, selected transmission schemes including a transmission scheme that regularly hops between precoding matrices, pilot symbol insertion scheme, IFFT (Inverse Fast Fourier Transform)/FFT, method of reducing PAPR, and guard interval insertion scheme) of each symbol group shown in
A frame configurator 7610 receives, as input, the baseband signal 7603 carrying PLPs, the baseband signal 7606 carrying P2 symbols, and the control signal 7609. On receipt of the input, the frame configurator 7610 changes the order of input data in frequency domain and time domain based on the information about frame structure included in the control signal, and outputs a (quadrature) baseband signal 7611_1 corresponding to stream 1 (a signal after the mapping, that is, a baseband signal based on a modulation scheme to be used) and a (quadrature) baseband signal 7611_2 corresponding to stream 2 (a signal after the mapping, that is, a baseband signal based on a modulation scheme to be used) both in accordance with the frame structure.
A signal processor 7612 receives, as input, the baseband signal 7611_1 corresponding to stream 1, the baseband signal 7611_2 corresponding to stream 2, and the control signal 7609 and outputs a modulated signal 1 (7613_1) and a modulated signal 2 (7613_2) each obtained as a result of signal processing based on the transmission scheme indicated by information included in the control signal 7609.
The characteristic feature noted here lies in the following. That is, when a transmission scheme that performs phase change on the signal after performing precoding (or after performing precoding, and switching the baseband signals) is selected, the signal processor performs phase change on signals after performing precoding (or after performing precoding, and switching the baseband signals) in a manner similar to
A pilot inserter 7614_1 receives, as input, the modulated signal 1 (7613_1) obtained as a result of the signal processing and the control signal 7609, inserts pilot symbols into the received modulated signal 1 (7613_1), and outputs a modulated signal 7615_1 obtained as a result of the pilot signal insertion. Note that the pilot symbol insertion is carried out based on information indicating the pilot symbol insertion scheme included the control signal 7609.
A pilot inserter 7614_2 receives, as input, the modulated signal 2 (7613_2) obtained as a result of the signal processing and the control signal 7609, inserts pilot symbols into the received modulated signal 2 (76132), and outputs a modulated signal 76152 obtained as a result of the pilot symbol insertion. Note that the pilot symbol insertion is carried out based on information indicating the pilot symbol insertion scheme included the control signal 7609.
An IFFT (Inverse Fast Fourier Transform) unit 7616_1 receives, as input, the modulated signal 7615_1 obtained as a result of the pilot symbol insertion and the control signal 7609, and applies IFFT based on the information about the IFFT method included in the control signal 7609, and outputs a signal 7617_1 obtained as a result of the IFFT.
An IFFT unit 7616_2 receives, as input, the modulated signal 7615_2 obtained as a result of the pilot symbol insertion and the control signal 7609, and applies IFFT based on the information about the IFFT method included in the control signal 7609, and outputs a signal 7617_2 obtained as a result of the IFFT.
A PAPR reducer 7618_1 receives, as input, the signal 7617_1 obtained as a result of the IFFT and the control signal 7609, performs processing to reduce PAPR on the received signal 7617_1, and outputs a signal 7619_1 obtained as a result of the PAPR reduction processing. Note that the PAPR reduction processing is performed based on the information about the PAPR reduction included in the control signal 7609.
A PAPR reducer 7618_2 receives, as input, the signal 7617_2 obtained as a result of the IFFT and the control signal 7609, performs processing to reduce PAPR on the received signal 7617_2, and outputs a signal 7619_2 obtained as a result of the PAPR reduction processing. Note that the PAPR reduction processing is carried out based on the information about the PAPR reduction included in the control signal 7609.
A guard interval inserter 7620_1 receives, as input, the signal 7619_1 obtained as a result of the PAPR reduction processing and the control signal 7609, inserts guard intervals into the received signal 7619_1, and outputs a signal 7621_1 obtained as a result of the guard interval insertion. Note that the guard interval insertion is carried out based on the information about the guard interval insertion scheme included in the control signal 7609.
A guard interval inserter 7620_2 receives, as input, the signal 7619_2 obtained as a result of the PAPR reduction processing and the control signal 7609, inserts guard intervals into the received signal 76192, and outputs a signal 7621_2 obtained as a result of the guard interval insertion. Note that the guard interval insertion is carried out based on the information about the guard interval insertion scheme included in the control signal 7609.
A P1 symbol inserter 7622 receives, as input, the signal 7621_1 obtained as a result of the guard interval insertion, the signal 7621_2 obtained as a result of the guard interval insertion, and the P1 symbol transmission data 7607, generates a P1 symbol signal from the P1 symbol transmission data 7607, adds the P1 symbol to the signal 7621_1 obtained as a result of the guard interval insertion, and adds the P1 symbol to the signal 7621_2 obtained as a result of the guard interval insertion. Then, the P1 symbol inserter 7622 outputs a signal 7623_1 as a result of the addition of the P1 symbol and a signal 7623_2 as a result of the addition of the P1 symbol. Note that a P1 symbol signal may be added to both the signals 7623_1 and 7623_2 or to one of the signals 7623_1 and 7623_2. In the case where the P1 symbol signal is added to one of the signals 7623_1 and 7623_2, the following is noted. For purposes of description, an interval of the signal to which a P1 symbol is added is referred to as a P1 symbol interval. Then, the signal to which a P1 signal is not added includes, as a baseband signal, a zero signal in an interval corresponding to the P1 symbol interval of the other signal.
A wireless processor 7624_1 receives the signal 7623_1 obtained as a result of the processing related to P1 symbol and the control signal 7609, performs processing such as frequency conversion, amplification, and the like, and outputs a transmission signal 7625_1. The transmission signal 7625_1 is then output as a radio wave from an antenna 7626_1.
A wireless processor 7624_2 receives the signal 7623_2 obtained as a result of the processing related to P1 symbol and the control signal 7609, performs processing such as frequency conversion, amplification, and the like, and outputs a transmission signal 7625_2. The transmission signal 7625_2 is then output as a radio wave from an antenna 7626_2.
As described above, by the P1 symbol, P2 symbol and control symbol group, information on transmission scheme of each PLP (for example, a transmission scheme of transmitting a single modulated signal, a transmission scheme of performing phase change on the signal after performing precoding (or after performing precoding, and switching the baseband signals)) and a modulation scheme being used is transmitted to a terminal. In this case, if the terminal extracts only PLP that is necessary as information to perform demodulation (including separation of signals and signal detection) and error correction decoding, power consumption of the terminal is reduced. Therefore, as described using
For example, assume that the broadcast station transmits each symbol having the frame structure as shown in
Note that, in the following description, as an example, assume that seven phase changing values are prepared in the transmission scheme of performing phase change on the signal after performing precoding (or after performing precoding, and switching the baseband signals), and are referred to as #0, #1, #2, #3, #4, #5 and #6. The phase changing values are to be regularly and cyclically used. That is to say, the phase changing values are to be regularly and cyclically changed in the order such as #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, #0, #1, #2, #3, #4, #5, #6, . . . .
As shown in
This is to say, in PLP $1, the first slot is the time T and the carrier 3, the second slot is the time T and the carrier 4, the third slot is the time T and a carrier 5, . . . , the seventh slot is a time T+1 and a carrier 1, the eighth slot is the time T+1 and a carrier 2, the ninth slot is the time T+1 and the carrier 3, . . . , the fourteenth slot is the time T+1 and a carrier 8, the fifteenth slot is a time T+2 and a carrier 0, . . . .
The slot (symbol) in PLP $K starts with a time S and a carrier 4 (7703 in
This is to say, in PLP $K, the first slot is the time S and the carrier 4, the second slot is the time S and a carrier 5, the third slot is the time S and a carrier 6, . . . , the fifth slot is the time S and a carrier 8, the ninth slot is a time S+1 and a carrier 1, the tenth slot is the time S+1 and a carrier 2, . . . , the sixteenth slot is the time S+1 and the carrier 8, the seventeenth slot is a time S+2 and a carrier 0, . . . .
Note that information on slot that includes information on the first slot (symbol) and the last slot (symbol) in each PLP and is used by each PLP is transmitted by the control symbol including the P1 symbol, the P2 symbol and the control symbol group.
In this case, as described using
Also, the first slot in another PLP transmitted using a transmission scheme that performs phase change on the signal after performing precoding (or after performing precoding, and switching the baseband signals) is precoded using the precoding matrix #0.
With the abovementioned scheme, an effect of suppressing the problems described in Embodiment D2 above, occurring in (a) and (b) is obtained.
Naturally, the reception device extracts necessary PLP from the information on slot that is included in the control symbol including the P1 symbol, the P2 symbol and the control symbol group and is used by each PLP to perform demodulation (including separation of signals and signal detection) and error correction decoding. The reception device learns a phase change rule of regularly performing phase change on the signal after performing precoding (or after performing precoding, and switching the baseband signals) in advance (when there are a plurality of rules, the transmission device transmits information on the rule to be used, and the reception device learns the rule being used by obtaining the transmitted information). By synchronizing a timing of rules of switching the phase changing values based on the number of the first slot in each PLP, the reception device can perform demodulation of information symbols (including separation of signals and signal detection).
Next, a case where the broadcast station (base station) transmits a modulated signal having a frame structure shown in
In
In this case, as described above, when the abovementioned transmission scheme for regularly performing phase change on the signal after performing precoding (or after performing precoding, and switching the baseband signals) is used in the subframe 7801, the first slot in PLP (PLP $1 (7802_1) through PLP $M (7802_M)) is assumed to be precoded using the precoding matrix #0 (referred to as initialization of the precoding matrices). The abovementioned initialization of precoding matrices, however, is irrelevant to a PLP in which another transmission scheme, for example, one of the transmission scheme not performing phase change, the transmission scheme using the spacetime block codes and the transmission scheme using a spatial multiplexing MIMO system (see
As shown in
In this case, the first slot (7701 in
Similarly, the first slot (7901 in
As described above, in each main frame, the first slot in the first PLP in the subframe for transmitting a plurality of modulated signals is characterized by being subject to phase change using the phase changing value #0.
This is also important to suppress the problems described in Embodiment D2 occurring in (a) and (b).
Note that since the first slot (7701 in
Similarly, note that since the first slot (7901 in
Note that, in the present embodiment, cases where (i) the transmission device in
The transmission devices pertaining to the present invention, as illustrated by
The schemes for regularly performing phase change on the modulated signal after precoding described in Embodiments 1 through 4, Embodiment A1, Embodiments C1 through C7, Embodiments D1 through D3 and Embodiment E1 are applicable to any baseband signals s1 and s2 mapped in the I (inphase)Q (quadrature(phase)) plane. Therefore, in Embodiments 1 through 4, Embodiment A1, Embodiments C1 through C7, Embodiments D1 through D3 and Embodiment E1, the baseband signals s1 and s2 have not been described in detail. On the other hand, when the scheme for regularly performing phase change on the modulated signal after precoding is applied to the baseband signals s1 and s2 generated from the error correction coded data, excellent reception quality can be achieved by controlling average power (average value) of the baseband signals s1 and s2. In the present embodiment, the following describes a scheme of setting the average power of s1 and s2 when the scheme for regularly performing phase change on the modulated signal after precoding is applied to the baseband signals s1 and s2 generated from the error correction coded data.
As an example, the modulation schemes for the baseband signal s1 and the baseband signal s2 are described as QPSK and 16QAM, respectively.
Since the modulation scheme for s1 is QPSK, s1 transmits two bits per symbol. Let the two bits to be transmitted be referred to as b0 and b1. On the other hand, since the modulation scheme for s2 is 16QAM, s2 transmits four bits per symbol. Let the four bits to be transmitted be referred to as b2, b3, b4 and b5. The transmission device transmits one slot composed of one symbol for s1 and one symbol for s2, i.e. six bits b0, b1, b2, b3, b4 and b5 per slot.
For example, in
Also, in
Here, assume that the average power of s1 is equal to the average power of s2, i.e. h shown in
[Math. 80]
√{square root over (2)}z (formula 80)
On the other hand, when h is represented by formula 78 in
A minimum Euclidian distance between signal points in the I (inphase)Q (quadrature(phase)) plane for 16QAM is as formula 81.
If the reception device performs error correction decoding (e.g. belief propagation decoding such as a sumproduct decoding in a case where the communication system uses LDPC codes) under this situation, due to a difference in reliability that “the absolute values of the loglikelihood ratio for b0 and b1 are higher than the absolute values of the loglikelihood ratio for b2 through b5”, a problem that the data reception quality degrades in the reception device by being affected by the absolute values of the loglikelihood ratio for b2 through b5 arises.
In order to overcome the problem, the difference between the absolute values of the loglikelihood ratio for b0 and b1 and the absolute values of the loglikelihood ratio for b2 through b5 should be reduced compared with
Therefore, it is considered that the average power (average value) of s1 is made to be different from the average power (average value) of s2.
The following explains some examples of operations of the power changer.
First, an example of the operation is described using
The power changer (8401B) receives a (mapped) baseband signal 307B for the modulation scheme 16QAM and a control signal (8400) as input. Letting a value for power change set based on the control signal (8400) be u, the power changer outputs a signal (8402B) obtained by multiplying the (mapped) baseband signal 307B for the modulation scheme 16QAM by u. Let u be a real number, and u>1.0. Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. e^{jθ(t)}, the following formula is satisfied.
Therefore, a ratio of the average power for QPSK to the average power for 16QAM is set to 1:u^{2}. With this structure, the reception device is in a reception condition in which the absolute value of the loglikelihood ratio shown in
The following describes a case where u in the ratio of the average power for QPSK to the average power for 16QAM 1:u^{2 }is set as shown in the following formula.
[Math. 83]
u=√{square root over (5)} (formula 83)
In this case, the minimum Euclidian distance between signal points in the I (inphase)Q (quadrature(phase)) plane for QPSK and the minimum Euclidian distance between signal points in the I (inphase)Q (quadrature(phase)) plane for 16QAM can be the same. Therefore, excellent reception quality can be achieved.
The condition that the minimum Euclidian distances between signal points in the I (inphase)Q (quadrature(phase)) plane for two different modulation schemes are equalized, however, is a mere example of the scheme of setting the ratio of the average power for QPSK to the average power for 16QAM. For example, according to other conditions such as a code length and a coding rate of an error correction code used for error correction codes, excellent reception quality may be achieved when the value u for power change is set to a value (higher value or lower value) different from the value at which the minimum Euclidian distances between signal points in the I (inphase)Q (quadrature(phase)) plane for two different modulation schemes are equalized. In order to increase the minimum distance between candidate signal points obtained at the time of reception, a scheme of setting the value u as shown in the following formula is considered, for example.
[Math. 84]
u=√{square root over (2)} (formula 84)
The value, however, is set appropriately according to conditions required as a system. This will be described later in detail.
In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point.
The above describes that the value u for power change is set based on the control signal (8400). The following describes setting of the value u for power change based on the control signal (8400) in order to improve data reception quality in the reception device in detail.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a block length (the number of bits constituting one coding (encoded) block, and is also referred to as the code length) for the error correction coding used to generate s1 and s2 when the transmission device supports a plurality of block lengths for the error correction codes.
Examples of the error correction codes include block codes such as turbo codes or duobinary turbo codes using tailbiting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two groups. The encoded data having been distributed to the two groups is modulated in the modulation scheme for s1 and in the modulation scheme for s2 to generate the (mapped) baseband signals s1(t) and s2(t).
The control signal (8400) is a signal indicating the selected block length for the error correction codes described above. The power changer (8401B) sets the value u for power change according to the control signal (8400).
The present invention is characterized in that the power changer (8401B) sets the value u for power change according to the selected block length indicated by the control signal (8400). Here, a value for power change set according to a block length X is referred to as u_{LX}.
For example, when 1000 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L1000}. When 1500 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L1500}. When 3000 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L3000}. In this case, for example, by setting u_{L1000}, u_{L1500 }and u_{L3000 }so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u_{L1000}=u_{L1500 }may be satisfied. What is important is that two or more values exist in u_{L1000}, u_{L1500 }and u_{L3000}).
Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the code length is set, and performs power change.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a coding rate for the error correction codes used to generate s1 and s2 when the transmission device supports a plurality of coding rates for the error correction codes.
Examples of the error correction codes include block codes such as turbo codes or duobinary turbo codes using tailbiting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two groups. The encoded data having been distributed to the two groups is modulated in the modulation scheme for s1 and in the modulation scheme for s2 to generate the (mapped) baseband signals s1(t) and s2(t).
The control signal (8400) is a signal indicating the selected coding rate for the error correction codes described above. The power changer (8401B) sets the value u for power change according to the control signal (8400).
The present invention is characterized in that the power changer (8401B) sets the value u for power change according to the selected coding rate indicated by the control signal (8400). Here, a value for power change set according to a coding rate rx is referred to as urX.
For example, when r1 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r1}. When r2 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r2}. When r3 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r3}. In this case, for example, by setting u_{r1}, u_{r2 }and u_{r3 }so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, u_{r1}=u_{r2 }may be satisfied. What is important is that two or more values exist in u_{r1}, u_{r2 }and ur3).
Note that, as examples of r1, r2 and r3 described above, coding rates 1/2, 2/3 and 3/4 are considered when the error correction code is the LDPC code.
Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the coding rate is set, and performs power change.
In order for the reception device to achieve excellent data reception quality, it is important to implement the following.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a modulation scheme used to generate s1 and s2 when the transmission device supports a plurality of modulation schemes.
Here, as an example, a case where the modulation scheme for s1 is fixed to QPSK and the modulation scheme for s2 is changed from 16QAM to 64QAM by the control signal (or can be set to either 16QAM or 64QAM) is considered. Note that, in a case where the modulation scheme for s2(t) is 64QAM, the mapping scheme for s2(t) is as shown in
By performing mapping in this way, the average power obtained when h in
That is to say, in
In
Note that, in the above description, the “modulation scheme for s1 is fixed to QPSK”. It is also considered that the modulation scheme for s2 is fixed to QPSK. In this case, power change is assumed to be not performed for the fixed modulation scheme (here, QPSK), and to be performed for a plurality of modulation schemes that can be set (here, 16QAM and 64QAM). That is to say, in this case, the transmission device does not have the structure shown in
When the modulation scheme for s2 is fixed to QPSK and the modulation scheme for s1 is changed from 16QAM to 64QAM (is set to either 16QAM or 64QAM), the relationship u_{16}<u_{64 }should be satisfied (note that a multiplied value for power change in 16QAM is u_{16}, a multiplied value for power change in 64QAM is u_{64}, and power change is not performed in QPSK).
Also, when a set of the modulation scheme for s1 and the modulation scheme for s2 can be set to any one of a set of QPSK and 16QAM, a set of 16QAM and QPSK, a set of QPSK and 64QAM and a set of 64QAM and QPSK, the relationship u_{16}<U_{64 }should be satisfied.
The following describes a case where the abovementioned description is generalized.
Let the modulation scheme for s1 be fixed to a modulation scheme C in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is c. Also, let the modulation scheme for s2 be set to either a modulation scheme A in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is a or a modulation scheme B in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is b (a>b>c) (however, let the average power (average value) for s2 in the modulation scheme A be equal to the average power (average value) for s2 in the modulation scheme B).
In this case, a value for power change set when the modulation scheme A is set to the modulation scheme for s2 is ua. Also, a value for power change set when the modulation scheme B is set to the modulation scheme for s2 is u_{b}. In this case, when the relationship u_{b}<u_{a }is satisfied, excellent data reception quality is obtained in the reception device.
Power change is assumed to be not performed for the fixed modulation scheme (here, modulation scheme C), and to be performed for a plurality of modulation schemes that can be set (here, modulation schemes A and B). When the modulation scheme for s2 is fixed to the modulation scheme C and the modulation scheme for s1 is changed from the modulation scheme A to the modulation scheme B (is set to either the modulation schemes A or B), the relationship u_{b}<u_{a }should be satisfied. Also, when a set of the modulation scheme for s1 and the modulation scheme for s2 can be set to any one of a set of the modulation scheme C and the modulation scheme A, a set of the modulation scheme A and the modulation scheme C, a set of the modulation scheme C and the modulation scheme B and a set of the modulation scheme B and the modulation scheme C, the relationship u_{b}<ua should be satisfied.
The following describes an example of the operation different from that described in Example 1, using
The power changer (8401B) receives a (mapped) baseband signal 307B for the modulation scheme 16QAM and a control signal (8400) as input. Letting a value for power change set based on the control signal (8400) be u, the power changer outputs a signal (8402B) obtained by multiplying the (mapped) baseband signal 307B for the modulation scheme 16QAM by u. Let u be a real number, and u<1.0. Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. e^{jθ(t)}, formula 82 is satisfied.
Therefore, a ratio of the average power for 64QAM to the average power for 16QAM is set to 1:u^{2}. With this structure, the reception device is in a reception condition as shown in
In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point.
The above describes that the value u for power change is set based on the control signal (8400). The following describes setting of the value u for power change based on the control signal (8400) in order to improve data reception quality in the reception device in detail.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a block length (the number of bits constituting one coding (encoded) block, and is also referred to as the code length) for the error correction codes used to generate s1 and s2 when the transmission device supports a plurality of block lengths for the error correction codes.
Examples of the error correction codes include block codes such as turbo codes or duobinary turbo codes using tailbiting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two groups. The encoded data having been distributed to the two groups is modulated in the modulation scheme for s1 and in the modulation scheme for s2 to generate the (mapped) baseband signals s1(t) and s2(t).
The control signal (8400) is a signal indicating the selected block length for the error correction codes described above. The power changer (8401B) sets the value u for power change according to the control signal (8400).
The present invention is characterized in that the power changer (8401B) sets the value u for power change according to the selected block length indicated by the control signal (8400). Here, a value for power change set according to a block length X is referred to as u_{LX}.
For example, when 1000 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L1000}. When 1500 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L1500}. When 3000 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L3000}. In this case, for example, by setting u_{L1000}, u_{L1500 }and u_{L3000 }so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u_{L1000}=u_{L1500 }may be satisfied. What is important is that two or more values exist in u_{L1000}, u_{L1500 }and u_{L3000}).
Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the code length is set, and performs power change.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a coding rate for the error correction codes used to generate s1 and s2 when the transmission device supports a plurality of coding rates for the error correction codes.
Examples of the error correction codes include block codes such as turbo codes or duobinary turbo codes using tailbiting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two groups. The encoded data having been distributed to the two groups is modulated in the modulation scheme for s1 and in the modulation scheme for s2 to generate the (mapped) baseband signals s1(t) and s2(t).
The control signal (8400) is a signal indicating the selected coding rate for the error correction codes described above. The power changer (8401B) sets the value u for power change according to the control signal (8400).
The present invention is characterized in that the power changer (8401B) sets the value u for power change according to the selected coding rate indicated by the control signal (8400). Here, a value for power change set according to a coding rate_{rx }is referred to as u_{rx}.
For example, when r1 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r1}. When r2 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r2}. When r3 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r3}. In this case, for example, by setting u_{r1}, u_{r2 }and u_{r3 }so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, u_{r1}=u_{r2 }may be satisfied. What is important is that two or more values exist in u_{r1}, u_{r2 }and u_{r3}).
Note that, as examples of r1, r2 and r3 described above, coding rates 1/2, 2/3 and 3/4 are considered when the error correction code is the LDPC code.
Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the coding rate is set, and performs power change.
In order for the reception device to achieve excellent data reception quality, it is important to implement the following.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a modulation scheme used to generate s1 and s2 when the transmission device supports a plurality of modulation schemes.
Here, as an example, a case where the modulation scheme for s1 is fixed to 64QAM and the modulation scheme for s2 is changed from 16QAM to QPSK by the control signal (or can be set to either 16QAM or QPSK) is considered. In a case where the modulation scheme for s1 is 64QAM, the mapping scheme for s1(t) is as shown in
By performing mapping in this way, the average power in 16QAM becomes equal to the average power (average value) in QPSK.
In
Note that, in the above description, the modulation scheme for s1 is fixed to 64QAM. When the modulation scheme for s2 is fixed to 64QAM and the modulation scheme for s1 is changed from 16QAM to QPSK (is set to either 16QAM or QPSK), the relationship u_{4}<u_{16 }should be satisfied (the same considerations should be made as the example 13) (note that a multiplied value for power change in 16QAM is u_{16}, a multiplied value for power change in QPSK is u_{4}, and power change is not performed in 64QAM). Also, when a set of the modulation scheme for s1 and the modulation scheme for s2 can be set to any one of a set of 64QAM and 16QAM, a set of 16QAM and 64QAM, a set of 64QAM and QPSK and a set of QPSK and 64QAM, the relationship u_{4}<u_{16 }should be satisfied.
The following describes a case where the abovementioned description is generalized.
Let the modulation scheme for s1 be fixed to a modulation scheme C in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is c. Also, let the modulation scheme for s2 be set to either a modulation scheme A in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is a or a modulation scheme B in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is b (c>b>a) (however, let the average power (average value) for s2 in the modulation scheme A be equal to the average power (average value) for s2 in the modulation scheme B).
In this case, a value for power change set when the modulation scheme A is set to the modulation scheme for s2 is ua. Also, a value for power change set when the modulation scheme B is set to the modulation scheme for s2 is u_{b}. In this case, when the relationship u_{a}<u_{b }is satisfied, excellent data reception quality is obtained in the reception device.
Power change is assumed to be not performed for the fixed modulation scheme (here, modulation scheme C), and to be performed for a plurality of modulation schemes that can be set (here, modulation schemes A and B). When the modulation scheme for s2 is fixed to the modulation scheme C and the modulation scheme for s1 is changed from the modulation scheme A to the modulation scheme B (is set to either the modulation schemes A or B), the relationship ua<u_{b }should be satisfied. Also, when a set of the modulation scheme for s1 and the modulation scheme for s2 can be set to any one of a set of the modulation scheme C and the modulation scheme A, a set of the modulation scheme A and the modulation scheme C, a set of the modulation scheme C and the modulation scheme B and a set of the modulation scheme B and the modulation scheme C, the relationship ua<u_{b }should be satisfied.
The following describes an example of the operation different from that described in Example 1, using
The power changer (8401B) receives a (mapped) baseband signal 307B for the modulation scheme 64QAM and a control signal (8400) as input. Letting a value for power change set based on the control signal (8400) be u, the power changer outputs a signal (8402B) obtained by multiplying the (mapped) baseband signal 307B for the modulation scheme 64QAM by u. Let u be a real number, and u>1.0. Letting the precoding matrix used in the scheme for regularly performing phase change on the modulated signal after precoding be F and the phase changing value used for regularly performing phase change be y(t) (y(t) may be imaginary number having the absolute value of 1, i.e. e^{jθ(t)}, formula 82 is satisfied.
Therefore, a ratio of the average power for 16QAM to the average power for 64QAM is set to 1:u^{2}. With this structure, the reception device is in a reception condition as shown in
In the conventional technology, transmission power control is generally performed based on feedback information from a communication partner. The present invention is characterized in that the transmission power is controlled regardless of the feedback information from the communication partner in the present embodiment. Detailed description is made on this point.
The above describes that the value u for power change is set based on the control signal (8400). The following describes setting of the value u for power change based on the control signal (8400) in order to improve data reception quality in the reception device in detail.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a block length (the number of bits constituting one coding (encoded) block, and is also referred to as the code length) for the error correction codes used to generate s1 and s2 when the transmission device supports a plurality of block lengths for the error correction codes.
Examples of the error correction codes include block codes such as turbo codes or duobinary turbo codes using tailbiting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of block lengths are supported. Encoded data for which error correction codes whose block length is selected from among the plurality of supported block lengths has been performed is distributed to two groups. The encoded data having been distributed to the two groups is modulated in the modulation scheme for s1 and in the modulation scheme for s2 to generate the (mapped) baseband signals s1(t) and s2(t).
The control signal (8400) is a signal indicating the selected block length for the error correction codes described above. The power changer (8401B) sets the value u for power change according to the control signal (8400).
The present invention is characterized in that the power changer (8401B) sets the value u for power change according to the selected block length indicated by the control signal (8400). Here, a value for power change set according to a block length X is referred to as u_{LX}.
For example, when 1000 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L1000}. When 1500 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L1500}. When 3000 is selected as the block length, the power changer (8401B) sets a value for power change to u_{L3000}. In this case, for example, by setting u_{L1000}, u_{L1500 }and u_{L3000 }so as to be different from one another, a high error correction capability can be achieved for each code length. Depending on the set code length, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the code length is changed, it is unnecessary to change the value for power change (for example, u_{L1000}=u_{L1500 }may be satisfied. What is important is that two or more values exist in u_{L1000}, u_{L1500 }and u_{L3000}).
Although the case of three code lengths is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more code lengths that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the code length is set, and performs power change.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a coding rate for the error correction codes used to generate s1 and s2 when the transmission device supports a plurality of coding rates for the error correction codes.
Examples of the error correction codes include block codes such as turbo codes or duobinary turbo codes using tailbiting, LDPC codes, or the like. In many communication systems and broadcasting systems, a plurality of coding rates are supported. Encoded data for which error correction codes whose coding rate is selected from among the plurality of supported coding rates has been performed is distributed to two groups. The encoded data having been distributed to the two groups is modulated in the modulation scheme for s1 and in the modulation scheme for s2 to generate the (mapped) baseband signals s1(t) and s2(t).
The control signal (8400) is a signal indicating the selected coding rate for the error correction codes described above. The power changer (8401B) sets the value u for power change according to the control signal (8400).
The present invention is characterized in that the power changer (8401B) sets the value u for power change according to the selected coding rate indicated by the control signal (8400). Here, a value for power change set according to a coding rate rx is referred to as ux.
For example, when r1 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r1}. When r2 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r2}. When r3 is selected as the coding rate, the power changer (8401B) sets a value for power change to u_{r3}. In this case, for example, by setting u_{r1}, u_{r2 }and u_{r3 }so as to be different from one another, a high error correction capability can be achieved for each coding rate. Depending on the set coding rate, however, the effect might not be obtained even if the value for power change is changed. In such a case, even when the coding rate is changed, it is unnecessary to change the value for power change (for example, u_{r1}=u_{r2 }may be satisfied. What is important is that two or more values exist in u_{r1}, u_{r2 }and u_{r3}).
Note that, as examples of r1, r2 and r3 described above, coding rates 1/2, 2/3 and 3/4 are considered when the error correction code is the LDPC code.
Although the case of three coding rates is taken as an example in the above description, the present invention is not limited to this. The important point is that two or more values for power change exist when there are two or more coding rates that can be set, and the transmission device selects any of the values for power change from among the two or more values for power change when the coding rate is set, and performs power change.
In order for the reception device to achieve excellent data reception quality, it is important to implement the following.
The following describes a scheme of setting the average power (average values) of s1 and s2 according to a modulation scheme used to generate s1 and s2 when the transmission device supports a plurality of modulation schemes.
Here, as an example, a case where the modulation scheme for s1 is fixed to 16QAM and the modulation scheme for s2 is changed from 64QAM to QPSK by the control signal (or can be set to either 64QAM or QPSK) is considered. In a case where the modulation scheme for s1 is 16QAM, the mapping scheme for s2(t) is as shown in
By performing mapping in this way, the average power in 16QAM becomes equal to the average power in QPSK.
In
Note that, in the above description, the modulation scheme for s1 is fixed to 16QAM. When the modulation scheme for s2 is fixed to 16QAM and the modulation scheme for s1 is changed from 64QAM to QPSK (is set to either 64QAM or QPSK), the relationship u_{4}<u_{64 }should be satisfied (the same considerations should be made as the example 13) (note that a multiplied value for power change in 64QAM is u_{64}, a multiplied value for power change in QPSK is u_{4}, and power change is not performed in 16QAM). Also, when a set of the modulation scheme for s1 and the modulation scheme for s2 can be set to any one of a set of 16QAM and 64QAM, a set of 64QAM and 16QAM, a set of 16QAM and QPSK and a set of QPSK and 16QAM, the relationship u_{4}<u_{64 }should be satisfied.
The following describes a case where the abovementioned description is generalized.
Let the modulation scheme for s1 be fixed to a modulation scheme C in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is c. Also, let the modulation scheme for s2 be set to either a modulation scheme A in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is a or a modulation scheme B in which the number of signal points in the I (inphase)Q (quadrature(phase)) plane is b (c>b>a) (however, let the average power (average value) for s2 in the modulation scheme A be equal to the average power (average value) for s2 in the modulation scheme B).
In this case, a value for power change set when the modulation scheme A is set to the modulation scheme for s2 is ua. Also, a value for power change set when the modulation scheme B is set to the modulation scheme for s2 is u_{b}. In this case, when the relationship u_{a}<u_{b }is satisfied, excellent data reception quality is obtained in the reception device.
Power change is assumed to be not performed for the fixed modulation scheme (here, modulation scheme C), and to be performed for a plurality of modulation schemes that can be set (here, modulation schemes A and B). When the modulation scheme for s2 is fixed to the modulation scheme C and the modulation scheme for s1 is changed from the modulation scheme A to the modulation scheme B (is set to either the modulation schemes A or B), the relationship ua<u_{b }should be satisfied. Also, when a set of the modulation scheme for s1 and the modulation scheme for s2 can be set to any one of a set of the modulation scheme C and the modulation scheme A, a set of the modulation scheme A and the modulation scheme C, a set of the modulation scheme C and the modulation scheme B and a set of the modulation scheme B and the modulation scheme C, the relationship ua<u_{b }should be satisfied.
The case where power change is performed for one of the modulation schemes for s1 and